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9 Removing Aux Winding Ringing to Resolve False Triggering of OVP and UVLO Faults

Parasitic inductances and capacitances are the major cause of aux winding ringing that can falsely trip OVP is in the design. To help reduce this ringing it is recommended through layout and transformer design that you keep the parasitic inductances and capacitances as small as possible.

The layout section (Section 10) of UCC28704 data sheet (SLUSCA8A) gives recommendations on how to layout the PSR flyback with minimal trace inductance and capacitance. It also has a layout that was constructed based on these recommendations in section 10.2, SLUSCA8.

When selecting and or designing your transformer (T1) it is recommended that a transformer have a primary leakage inductance (L_PLK) of less than three percent of the primary magnetizing inductance (L_PM), Figure 1-1. This will help to reduce ringing at the switch nodes.

Equation 12.

L_{P L K} \leq 0.03 \times L_{P M}

During layout process, keep the PCB traces in the power stage; as short as, possible. Keep in mind that every inch of trace adds roughly 10 nH of parasitic trace inductance (L_TRACE). Keeping the traces as short as possible removes unwanted antennas from the design helping to improve noise impunity as well..

Equation 13.

L_{T R A C E} ≅ \frac{10 n H}{i n} = \frac{10 n H}{2.54 c m}

Use an RCD clamp (R_A , R_C, C_A, D_A) over a TVS clamp (D_A, D_Z), (Figure 1-1). An RCD clamp will provide dampening at the switch node, where TVS clamp will only clamp the voltage when the switch node rings above the clamp voltage and provides very little to no dampening. Set the R_A and C_A time constant of the clamp to greater than 10 times the maximum switching period, Equation 14 and Equation 15. Please note for safety the designer want to use D_Z to clamp V_SW as well.

Equation 14.

R_{A} \times C_{A} \geq \frac{10}{f_{S W (m a x)}} = \frac{10}{100 k H z} = 100 µ s

Equation 15.

R_{A} \times C_{A} = 511 k Ω \times 1 n F = 511 µ s

To reduce excessive ringing across the secondary (V_SEC ) winding it will couple into V_AUX through the auxiliary to primary turns ratio. The waveform presented in Figure 9-1 shows ringing on the secondary of simulated from a 390 V to 12 V, 10 W, flyback converter. Excessive ringing present on V_SEC will couple onto V_AUX and if it severe enough can cause and OVP. As a result, the design will not lose output voltage regulation causing the design to misbehave.

Figure 9-1 Flyback Secondary Voltage with Excessive Ringing

The cycle by cycle energy transfer between parasitic leakage inductances and parasitic switch node capacitances causes ringing at the flyback converter switch nodes. This ringing will couple through the flyback converter’s switch nodes to V_AUX. Excessive voltage ringing on V_AUX can accidentally trigger an OVP.

The energy cycling by the transformer’s primary leakage inductance (L_PLK) and primary switch node (C_SW1) is one contributor this high frequency V_AUX ringing that causes OVP issues. Another contributor is the energy cycling by the transformer’s secondary leakage inductance (L_SLK) and the secondary switch node capacitance (C_SW2). This excessive ringing can generally be dampened with an RC snubber (R_B, R_C) across the converters output rectifier (D_C) shown in Figure 9-2.

Figure 9-2 Simplified Flyback Schematic with Parasitic.

To give an example of how to implement a snubber circuit, a 12 V, 10 W flyback design was created and simulated. The waveforms from this circuit are presented in Figure 9-2 and would trigger an OVP fault incorrectly due to ringing on the secondary winding (V_SEC) through the transformer to V_AUX.

Figure 9-3 Measuring Low Frequency Ringing to Calculate C_SW2

To setup the snubber requires knowing or calculating the transformer’s primary ( L_PM) and secondary L_SM magnetizing induct, L_SLK, C_SW2. With this information the secondary magnetizing inductance (L_SM) can be calculated by knowing the transformer primary to secondary turns ratio (N_P/N_S) and primary magnetizing inductance (L_PM) which are given in the transformer data sheet and using equations Equation 16 and Equation 17. For this example the transformer had an L_PM of 680 µH and N_P/N_S of 5.8. L_SM for this design was calculated to be 20 µH.

Equation 16.

\frac{N_{P}}{N_{A}} = 5.8

Equation 17.

L_{S M} = \frac{L_{P M}}{{(\frac{N_{P}}{N_{S}})}^{2}} = \frac{680 µ H}{{5.8}^{2}} ≅ 20 µ H

To calculate C_SW2 requires knowing L_SM and studying the V_SEC waveform and measuring the low frequency ringing (f_r1) during the t_D time interval, Figure 9-3. f_r1 should be measured when the flyback converter is operating at light load and operating deep into DCM. In this example f_r1 was measured to be 645 kHz. C_SW2 can then be calculated using Equation 19, which for this example was 3 nF

Equation 18.

f_{r 1} = 645 k H z

Equation 19.

C_{S W 2} = \frac{1}{{(2 \times π \times f_{r 1})}^{2} \times L_{S M}} = \frac{1}{{(2 \times π \times 645 k H z)}^{2} \times 20 µ H} ≅ 3 n F

The next step is to measure the high frequency ringing (f_r2). during time interval t_DMAG, Figure 9-4. This resonant frequency is caused by the interaction of C_SW2 and L_SECP. Based on f_r2and C_SW2, L_SECP can be calculated using Equation 21. With a measured f_r2 of 14 MHz and C_SW2 of 3 nF the calculated L_SECP is approximately 43 nH.

Figure 9-4 Measure High Frequency Ringing During Time Interval t_DMAG

Equation 20.

f_{r 2} = 14 M H z

,Measured high-freqeunecy ringing durring interval t_DMAG

Equation 21.

L_{S E C P} = \frac{1}{{(2 \times π \times f_{r 2})}^{2} \times C_{S W 2}} = \frac{1}{{(2 \times π \times 14 M h z)}^{2} \times 3 n F} \approx 43 n H

Snubbing resistor R_B is chosen to critically dampen the high-frequency ringing and can be calculated using Equation 22.

Equation 22.

R_{B} = \frac{1}{Q} \sqrt{\frac{L_{S E C P}}{C_{S w 2}}} = \frac{1}{1} \sqrt{\frac{43 n H}{3 n F}} ≅ 3.8 Ω

A standard resistor was chosen for resistor R_B :

R_B = 3.83

The snubbing capacitor C_C was chosen based on Equation 23, based on the converter's maximum nominal switching frequency (f_SW). By setting C_C this way the snubber will only be active for 1% of the switching period, keeping snubber losses to a minimum. The flyback design being evaluated had a f_SW of 75 kHz.

Equation 23.

C_{C} = \frac{0.01}{f_{S W} \times R_{B} \times 5} = \frac{0.01}{75 k H z \times 3.8 Ω \times 5} ≅ 7 n F

A standard capacitance value for C_C was chosen for the design:

C_C = 6.8 nF

The snubber components that were selected for R_B and R_C were applied to the circuit presented in Figure 1-1 and Figure 9-2. The result was the secondary winding was critically damped. Please refer to Figure 9-5 for damped waveforms results.

Figure 9-5 Snubber, RB 3.83 ohms, CC = 6.8nF