SLUAAC5 August 2021 UCC28700 , UCC28701 , UCC28702 , UCC28703 , UCC28704 , UCC28710 , UCC28711 , UCC28712 , UCC28713 , UCC28720 , UCC28722 , UCC28730 , UCC28740 , UCC28742 , UCC28910 , UCC28911

- Trademarks
- 1 Introduction
- 2 Brief Review of DCM FM, AM, FM Flyback Control Law
- 3 Input (VIN) and Output (VOUT) Voltage Sensing for UVLO and OVP Fault Protection
- 4 Input Under Voltage Lockout (UVLO) Protection
- 5 Output Overvoltage (OVP) Protection
- 6 Not Recognizing a UVLO or OVP Fault
- 7 Separate Bias Supply Startup Issue and Resolution
- 8 Not Having a Clean Aux Winding Signal
- 9 Removing Aux Winding Ringing to Resolve False Triggering of OVP and UVLO Faults
- 10Noise on CS Pin Tripping Over Current Protection (OCP)
- 11Summary
- 12References

Parasitic inductances and capacitances are the major cause of aux winding ringing that can falsely trip OVP is in the design. To help reduce this ringing it is recommended through layout and transformer design that you keep the parasitic inductances and capacitances as small as possible.

The layout section (Section 10) of UCC28704 data sheet (SLUSCA8A) gives recommendations on how to layout the PSR flyback with minimal trace inductance and capacitance. It also has a layout that was constructed based on these recommendations in section 10.2, SLUSCA8.

When selecting and or designing your
transformer (T1) it is recommended that a transformer have a primary leakage
inductance (L_{PLK}) of less than three percent of the primary magnetizing
inductance (L_{PM}), Figure 1-1. This will help to reduce ringing at the switch nodes.

Equation 12. ${L}_{PLK}\le 0.03\times {L}_{PM}$

During layout process, keep the PCB
traces in the power stage; as short as, possible. Keep in mind that every inch of
trace adds roughly 10 nH of parasitic trace inductance (L_{TRACE}). Keeping
the traces as short as possible removes unwanted antennas from the design helping to
improve noise impunity as well..

Equation 13. ${L}_{TRACE}\cong \frac{10nH}{in}=\frac{10nH}{2.54cm}$

Use an RCD clamp (R_{A} ,
R_{C}, C_{A}, D_{A}) over a TVS clamp (D_{A},
D_{Z}), (Figure 1-1). An RCD clamp will provide dampening at the switch node, where TVS clamp will
only clamp the voltage when the switch node rings above the clamp voltage and
provides very little to no dampening. Set the R_{A} and C_{A} time
constant of the clamp to greater than 10 times the maximum switching period, Equation 14 and Equation 15. Please note for safety the designer want to use D_{Z} to clamp
V_{SW} as well.

Equation 14. ${R}_{A}\times {C}_{A}\ge \frac{10}{{f}_{SW\left(max\right)}}=\frac{10}{100kHz}=100\mu s$

Equation 15. ${R}_{A}\times {C}_{A}=511k\Omega \times 1nF=511\mu s$

To reduce excessive ringing across the
secondary (V_{SEC} ) winding it will couple into V_{AUX} through the
auxiliary to primary turns ratio. The waveform presented in Figure 9-1 shows
ringing on the secondary of simulated from a 390 V to 12 V, 10 W, flyback converter.
Excessive ringing present on V_{SEC} will couple onto V_{AUX} and if
it severe enough can cause and OVP. As a result, the design will not lose output
voltage regulation causing the design to misbehave.

The cycle by cycle energy transfer
between parasitic leakage inductances and parasitic switch node capacitances causes
ringing at the flyback converter switch nodes. This ringing will couple through the
flyback converter’s switch nodes to V_{AUX}. Excessive voltage ringing on
V_{AUX} can accidentally trigger an OVP.

The energy cycling by the
transformer’s primary leakage inductance (L_{PLK}) and primary switch node
(C_{SW1}) is one contributor this high frequency V_{AUX} ringing
that causes OVP issues. Another contributor is the energy cycling by the
transformer’s secondary leakage inductance (L_{SLK}) and the secondary
switch node capacitance (C_{SW2}). This excessive ringing can generally be
dampened with an RC snubber (R_{B}, R_{C}) across the converters
output rectifier (D_{C}) shown in Figure 9-2.

To give an example of how to implement
a snubber circuit, a 12 V, 10 W flyback design was created and simulated. The
waveforms from this circuit are presented in Figure 9-2 and would trigger an OVP fault incorrectly due to ringing on the secondary
winding (V_{SEC}) through the transformer to V_{AUX}.

To setup the snubber requires knowing
or calculating the transformer’s primary ( L_{PM}) and secondary
L_{SM} magnetizing induct, L_{SLK}, C_{SW2}. With this
information the secondary magnetizing inductance (L_{SM}) can be calculated
by knowing the transformer primary to secondary turns ratio
(N_{P}/N_{S}) and primary magnetizing inductance
(L_{PM}) which are given in the transformer data sheet and using equations
Equation 16 and Equation 17. For this example the transformer had an L_{PM} of 680 µH and
N_{P}/N_{S} of 5.8. L_{SM} for this design was
calculated to be 20 µH.

Equation 16.
$\frac{{N}_{P}}{{N}_{A}}=5.8$

Equation 17.
${L}_{SM}=\frac{{L}_{PM}}{{\left(\frac{{N}_{P}}{{N}_{S}}\right)}^{2}}=\frac{680\mu H}{{5.8}^{2}}\cong 20\mu H$

To calculate C_{SW2} requires
knowing L_{SM} and studying the V_{SEC} waveform and measuring the
low frequency ringing (f_{r1}) during the t_{D} time interval, Figure 9-3. f_{r1} should be measured when the flyback converter is operating at
light load and operating deep into DCM. In this example f_{r1} was measured
to be 645 kHz. C_{SW2} can then be calculated using Equation 19, which for this example was 3 nF

Equation 18.
${f}_{r1}=645kHz$

Equation 19.
${C}_{SW2}=\frac{1}{{\left(2\times \pi \times {f}_{r1}\right)}^{2}\times {L}_{SM}}=\frac{1}{{\left(2\times \pi \times 645kHz\right)}^{2}\times 20\mu H}\cong 3nF$

The next step is to measure the high
frequency ringing (f_{r2}). during time interval t_{DMAG}, Figure 9-4. This resonant frequency is caused by the interaction of C_{SW2} and
L_{SECP}. Based on f_{r2 }and C_{SW2}, L_{SECP}
can be calculated using Equation 21. With a measured f_{r2} of 14 MHz and C_{SW2} of 3 nF the
calculated L_{SECP} is approximately 43 nH.

Equation 20.
${f}_{r2}=14MHz$
,Measured high-freqeunecy ringing durring interval
t_{DMAG}

Equation 21. ${L}_{SECP}=\frac{1}{{\left(2\times \pi \times {f}_{r2}\right)}^{2}\times {C}_{SW2}}=\frac{1}{{\left(2\times \pi \times 14Mhz\right)}^{2}\times 3nF}\approx 43nH$

Snubbing resistor R_{B} is
chosen to critically dampen the high-frequency ringing and can be calculated using
Equation 22.

Equation 22.
${R}_{B}=\frac{1}{Q}\sqrt{\frac{{L}_{SECP}}{{C}_{Sw2}}}=\frac{1}{1}\sqrt{\frac{43nH}{3nF}}\cong 3.8\Omega $

A standard resistor was chosen for
resistor R_{B} :

R_{B} = 3.83

The snubbing capacitor C_{C}
was chosen based on Equation 23, based on the converter's maximum nominal switching frequency (f_{SW}).
By setting C_{C} this way the snubber will only be active for 1% of the
switching period, keeping snubber losses to a minimum. The flyback design being
evaluated had a f_{SW} of 75 kHz.

Equation 23.
${C}_{C}=\frac{0.01}{{f}_{SW}\times {R}_{B}\times 5}=\frac{0.01}{75kHz\times 3.8\Omega \times 5}\cong 7nF$

A standard capacitance value for
C_{C} was chosen for the design:

C_{C} = 6.8 nF

The snubber components that were
selected for R_{B} and R_{C} were applied to the circuit presented
in Figure 1-1 and Figure 9-2. The result was the secondary winding was critically damped. Please refer to
Figure 9-5 for damped waveforms results.