ZHCSAO9A December   2012  – September 2015 TPS92690

PRODUCTION DATA.  

  1. 1 特性
  2. 2 应用
  3. 3 说明
  4. 4 修订历史记录
  5. 5 Pin Configuration and Functions
  6. 6 Specifications
    1. 6.1 Absolute Maximum Ratings
    2. 6.2 ESD Ratings
    3. 6.3 Recommended Operating Conditions
    4. 6.4 Thermal Information
    5. 6.5 Electrical Characteristics
    6. 6.6 Typical Characteristics
  7. 7 Detailed Description
    1. 7.1 Overview
    2. 7.2 Functional Block Diagram
    3. 7.3 Feature Description
      1. 7.3.1  Current Regulators
      2. 7.3.2  Peak Current Mode Control
      3. 7.3.3  Switching Frequency and Synchronization
      4. 7.3.4  Current Sense and Current Limit
      5. 7.3.5  Average LED Current
      6. 7.3.6  Precision Reference (VREF)
      7. 7.3.7  Low-Level Analog Dimming
      8. 7.3.8  Soft-Start and Shutdown
      9. 7.3.9  VCC Regulator and Start-Up
      10. 7.3.10 Overvoltage Protection (OVP)
      11. 7.3.11 Input Undervoltage Lockout (UVLO)
      12. 7.3.12 PWM Dimming
      13. 7.3.13 Control Loop Compensation
      14. 7.3.14 Thermal Shutdown
    4. 7.4 Device Functional Modes
  8. 8 Application and Implementation
    1. 8.1 Application Information
      1. 8.1.1 Inductor
      2. 8.1.2 LED Dynamic Resistance
      3. 8.1.3 Output Capacitor
      4. 8.1.4 Input Capacitor
      5. 8.1.5 MOSFET Selection
      6. 8.1.6 Recirculating Diode
    2. 8.2 Typical Applications
      1. 8.2.1 Basic Topology Schematics
        1. 8.2.1.1 Design Requirements
        2. 8.2.1.2 Detailed Design Procedure
          1. 8.2.1.2.1  Operating Point
          2. 8.2.1.2.2  Switching Frequency
          3. 8.2.1.2.3  Average LED Current
          4. 8.2.1.2.4  Inductor Ripple Current
          5. 8.2.1.2.5  Output Capacitance
          6. 8.2.1.2.6  Peak Current Limit
          7. 8.2.1.2.7  Loop Compensation
          8. 8.2.1.2.8  Input Capacitance
          9. 8.2.1.2.9  NFET
          10. 8.2.1.2.10 Diode
          11. 8.2.1.2.11 Input UVLO
          12. 8.2.1.2.12 Output OVLO
        3. 8.2.1.3 Application Curve
      2. 8.2.2 Simplified Application
        1. 8.2.2.1 Design Requirements
        2. 8.2.2.2 Detailed Design Procedure
          1. 8.2.2.2.1  Operating Point
          2. 8.2.2.2.2  Switching Frequency
          3. 8.2.2.2.3  Average LED Current
          4. 8.2.2.2.4  Inductor Ripple Current
            1. 8.2.2.2.4.1 Minimum Inductor Value
            2. 8.2.2.2.4.2 Inductor Ripple Current
            3. 8.2.2.2.4.3 RMS Inductor Current
          5. 8.2.2.2.5  LED Ripple Current
            1. 8.2.2.2.5.1 Output Capacitor
            2. 8.2.2.2.5.2 Output Capacitor RMS Current
          6. 8.2.2.2.6  Peak Current Limit
          7. 8.2.2.2.7  Loop Compensation
            1. 8.2.2.2.7.1 Compensation Capacitor
            2. 8.2.2.2.7.2 RHP Zero
            3. 8.2.2.2.7.3 Output Capacitor Pole
          8. 8.2.2.2.8  Input Capacitance
          9. 8.2.2.2.9  NFET
            1. 8.2.2.2.9.1 Maximum Average NFET Current
            2. 8.2.2.2.9.2 RMS Transistor Current
          10. 8.2.2.2.10 Diode
            1. 8.2.2.2.10.1 Maximum Average Diode Current
          11. 8.2.2.2.11 Output OVLO
          12. 8.2.2.2.12 Input UVLO
          13. 8.2.2.2.13 Soft-Start
          14. 8.2.2.2.14 PWM Dimming Method
          15. 8.2.2.2.15 Analog Dimming Method
  9. 9 Power Supply Recommendations
    1. 9.1 Bench Supply Current Limit
  10. 10Layout
    1. 10.1 Layout Guidelines
    2. 10.2 Layout Example
  11. 11器件和文档支持
    1. 11.1 社区资源
    2. 11.2 商标
    3. 11.3 静电放电警告
    4. 11.4 Glossary
  12. 12机械、封装和可订购信息

封装选项

机械数据 (封装 | 引脚)
散热焊盘机械数据 (封装 | 引脚)
订购信息

8 Application and Implementation

NOTE

Information in the following applications sections is not included in the TI component specification, and TI does not warrant its accuracy or completeness. TI customers are responsible for determining suitability of components for their purposes. Customers should validate and test their design implementation to confirm system functionality.

8.1 Application Information

8.1.1 Inductor

The inductor (L1) is the main energy storage device in a switching regulator. Depending on the topology, energy is stored in the inductor and transferred to the load in different ways (as an example, boost operation is detailed in the Current Regulators section). The size of the inductor, the voltage across it, and the length of the switching subinterval (tON or tOFF) determines the inductor current ripple (ΔiL-PP). In the design process, L1 is chosen to provide a desired ΔiL-PP. For a Cuk regulator the second inductor (L2) has a direct connection to the load, which is good for a current regulator. This requires little to no output capacitance therefore ΔiL-PP is basically equal to the LED ripple current ΔiLED-PP since the inductor ripple in L2 is equal to that in L1. However, for boost and other buck-boost regulators, there is always an output capacitor which reduces ΔiLED-PP, therefore the inductor ripple can be larger than in the Cuk regulator case where output capacitance is minimal or completely absent.

In general, ΔiLED-PP is recommended by manufacturers to be less than 40% of the average LED current (ILED). Therefore, for the Cuk regulator with no output capacitance, ΔiLED-PP should also be less than 40% of ILED unless a large output capacitor is used. For the boost and other buck-boost topologies, ΔiL-PP can be much higher depending on the output capacitance value. However, ΔiL-PP is suggested to be less than 100% of the average inductor current (iL) to limit the RMS inductor current. ΔiL-PP is defined as:

Equation 27. TPS92690 eq_delta_vsbk3.gif

Be sure to observe the minimum inductor value from the Control Loop Compensation section. L1 is also suggested to have an RMS current rating at least 25% higher than the calculated minimum allowable RMS inductor current (IL-RMS).

8.1.2 LED Dynamic Resistance

When the load is a string of LEDs, the output load resistance is the LED string dynamic resistance plus RCS. LEDs are PN junction diodes, and their dynamic resistance shifts as their forward current changes. Dividing the forward voltage of a single LED (VLED) by the forward current (ILED) can lead to an incorrect calculation of the dynamic resistance of a single LED (rLED). The result can be 5 to 10 times higher than the true rLED value.

TPS92690 dyn_resist_lvsbk3.gif Figure 18. Dynamic Resistance

Obtaining rLED is accomplished by referring to the manufacturer LED I-V characteristic. It can be calculated as the slope at the nominal operating point as shown in Figure 18. For any application with more than 2 series LEDs, RCS can be neglected allowing rD to be approximated as the number of LEDs multiplied by rLED.

8.1.3 Output Capacitor

For boost, SEPIC, and flyback regulators, the output capacitor (CO) provides energy to the load when the recirculating diode (D1) is reverse biased during the first switching subinterval. An output capacitor in a Cuk topology simply reduces the LED current ripple (ΔiLED-PP) below the inductor current ripple (ΔiL-PP). In all cases, CO is sized to provide a desired ΔiLED-PP. As mentioned in Inductor, ΔiLED-PP is recommended by manufacturers to be <40% of the average LED current (ILED).

CO should be carefully chosen to account for derating due to temperature and operating voltage. It must also have the necessary RMS current rating. Ceramic capacitors are the best choice due to their high ripple current rating, long lifetime, and good temperature performance. An X7R dielectric rating is suggested.

8.1.4 Input Capacitor

The input capacitor (CIN) only needs to provide the ripple current due to the direct connection to the inductor. CIN is selected given the maximum input voltage ripple (ΔVIN-PP) which can be tolerated. ΔVIN-PP is suggested to be less than 10% of the input voltage (VIN). An input capacitance at least 100% greater than the calculated CIN value is recommended to account for derating due to temperature and operating voltage. When PWM dimming, even more capacitance can be helpful to minimize the large current draw from the input voltage source during the rising transition of the LED current waveform.

The chosen input capacitors must also have the necessary RMS current rating. Ceramic capacitors are again the best choice due to their high ripple current rating, long lifetime, and good temperature performance. An X7R dielectric rating is suggested.

For most applications, it is recommended to bypass the VIN pin with an 0.1-µF ceramic capacitor placed as close as possible to the pin. In situations where the bulk input capacitance may be far from the TPS92690 device, a 10-Ω series resistor can be placed between the bulk input capacitance and the bypass capacitor, creating a 150-kHz filter to eliminate undesired high frequency noise.

8.1.5 MOSFET Selection

The TPS92690 device requires an external N-channel FET (Q1) as the main power MOSFET for the switching regulator. Q1 is recommended to have a voltage rating at least 15% higher than the maximum transistor voltage to ensure safe operation during the ringing of the switch node. In practice, all switching regulators have some ringing at the switch node due to the diode parasitic capacitance and the lead inductance. The current rating is recommended to be at least 10% higher than the average transistor current. The power rating is then verified by calculating the power loss given the average transistor current and the N-channel FET on-resistance (RDS(on)).

In general, the N-channel FET should be chosen to minimize total gate charge (Qg) when ƒSW is high and minimize RDS(on) otherwise. This minimizes the dominant power losses in the system. Frequently, higher current N-channel FETs in larger packages are chosen for better thermal performance.

8.1.6 Recirculating Diode

A recirculating diode (D1) is required to carry the inductor current during tOFF. The most efficient choice for D1 is a Schottky diode due to low forward voltage drop and near-zero reverse recovery time. Similar to Q1, D1 is recommended to have a voltage rating at least 15% higher than the maximum transistor voltage to ensure safe operation during the ringing of the switch node and a current rating at least 10% higher than the average diode current. The power rating is verified by calculating the power loss through the diode. This is accomplished by checking the typical diode forward voltage from the I-V curve on the product datasheet and multiplying by the average diode current. In general, higher current diodes have a lower forward voltage and come in better performing packages minimizing both power losses and temperature rise.

8.2 Typical Applications

8.2.1 Basic Topology Schematics

TPS92690 cuk_topo_buckb_cir_lvsbk3.gif Figure 19. CUK Topology (Buck-Boost)
TPS92690 quas_res_flyback_topo_cir_lvsbk3.gif Figure 20. Quasi-Resonant Flyback Topology
TPS92690 sepic_topo_buckb_cir_lvsbk3.gif Figure 21. SEPIC Topology (Buck-Boost)
TPS92690 boost_topo_pwm_dimm_cir_lvsbk3.gif Figure 22. Boost Topology With PWM Dimming

8.2.1.1 Design Requirements

N = 10

VLED = 3.5 V

rLED = 500 mΩ

VIN = 12 V

VIN-MIN = 8 V

VIN-MAX = 19 V

fSW = 420 kHz

VCS = 50 mV

ILED = 500 mA

ΔiL-PP < 650 mA

ΔiLED-PP < 50 mA

ΔvIN-PP = 50 mV

VLIM = 100 mV

ILIM = 5 A

VTURN-ON = 7.8 V

VHYS = 2 V

VTURN-OFF = 40 V

VHYSO = 5 V

8.2.1.2 Detailed Design Procedure

8.2.1.2.1 Operating Point

Solve for VO and rD:

Equation 28. TPS92690 Eq_slvsbk3_Vocalc.gif
Equation 29. TPS92690 Eq_slvsbk3_rDcalc.gif

Solve for D, DMAX, and DMIN:

Equation 30. TPS92690 Eq_slvsbk3_Dclac.gif
Equation 31. TPS92690 Eq_slvsbk3_Dminclac.gif
Equation 32. TPS92690 Eq_slvsbk3_Dmaxclac.gif

8.2.1.2.2 Switching Frequency

Solve for RT:

Equation 33. TPS92690 Eq_slvsbk3_RTcalc.gif

A close standard resistor is 105 kΩ resulting in fSW = 402 kHz. Choose RT = 105 kΩ.

8.2.1.2.3 Average LED Current

Solve for RCS using our desired 50-mV sense voltage:

Equation 34. TPS92690 Eq_slvsbk3_Rcscalc.gif

Solve for VIADJ:

Equation 35. TPS92690 Eq_slvsbk3_Vadjcalc.gif

A resistor divider can be used from the reference pin (VREF) to IADJ, select RIAD2 = 100 kΩ and solve for RIAD1:

Equation 36. TPS92690 Eq_slvsbk3_Radjcalc.gif

The closest standard value is to choose RADJ1 = 25.5 kΩ.

8.2.1.2.4 Inductor Ripple Current

Solve for the minimum value of L1 for stability:

Equation 37. TPS92690 Eq_slvsbk3_Lmincalc.gif

The inductor value required to meet the ripple current requirements is:

Equation 38. TPS92690 Eq_slvsbk3_iLripplecalc.gif

The closest standard inductor is 33 µH therefore ΔiL-PP is:

Equation 39. TPS92690 Eq_slvsbk3_iLripplecalc2.gif

Determine minimum allowable RMS current rating:

Equation 40. TPS92690 Eq_slvsbk3_iLrmscalc.gif

The chosen component is L1 = 33 µH.

8.2.1.2.5 Output Capacitance

Solve for CO:

Equation 41. TPS92690 Eq_slvsbk3_Coclac.gif

Add some capacitance to account for voltage de-rating and temperature and choose CO = 4.7 µF.

Determine minimum allowable RMS current rating:

Equation 42. TPS92690 Eq_slvsbk3_Cormscalc.gif

Since this is a PWM dimming application the output capacitor should be placed directly across the LED string and not connected to ground. So the CS pin should have additional filtering in the form of RF = 47 Ω and CF = 47 nF.

8.2.1.2.6 Peak Current Limit

Solve for RLIM:

Equation 43. TPS92690 Eq_slvsbk3_Rlimcalc.gif

The closest standard resistor is 0.02 Ω; therefore, choose RLIM = 0.02 Ω.

Assume RLIM2 = 100 kΩ and calculate RLIM1:

Equation 44. TPS92690 Eq_slvsbk3_Rlim1calc.gif

The closest standard value is RLIM1 = 4.22 kΩ

8.2.1.2.7 Loop Compensation

Check the frequency of the output pole:

Equation 45. TPS92690 Eq_slvsbk3_fpcocalc.gif

Check the frequency of the RHP zero:

Equation 46. TPS92690 Eq_slvsbk3_frhpzcalc.gif

The lower of the two is the RHP zero at 1.64 kHz, so the maximum crossover frequency should be 164 Hz or less. Calculate the minimum COMP capacitor value:

Equation 47. TPS92690 Eq_slvsbk3_Ccmpcalc.gif

To ensure stability over all conditions add some margin and choose CCMP = 47 nF.

8.2.1.2.8 Input Capacitance

Solve for the minimum CIN:

Equation 48. TPS92690 Eq_slvsbk3_Cinclac.gif

To minimize power supply interaction a 200% larger capacitance or more should be used particularly with PWM dimming, therefore the actual ΔvIN-PP is much lower. Choose CIN = 10 µF.

Determine minimum allowable RMS current rating:

Equation 49. TPS92690 Eq_slvsbk3_Cinrmscalc.gif

8.2.1.2.9 NFET

Determine minimum Q1 voltage rating and current rating:

Equation 50. TPS92690 Eq_slvsbk3_Vtmaxcalc.gif
Equation 51. TPS92690 Eq_slvsbk3_Itmaxcalc.gif

The RMS current rating used in conjunction with the chosen FET RDS-ON to calculate power dissipation is:

Equation 52. TPS92690 Eq_slvsbk3_Itrmscalc.gif

8.2.1.2.10 Diode

Determine minimum D1 voltage rating and current rating:

Equation 53. TPS92690 Eq_slvsbk3_Vrdmaxcalc.gif
Equation 54. TPS92690 Eq_slvsbk3_Idmaxcalc.gif

8.2.1.2.11 Input UVLO

Since this is a PWM dimming application RUVH will be used. Start by picking RUV2 = 10 kΩ and solve for RUV1:

Equation 55. TPS92690 Eq_slvsbk3_Ruv1calc.gif

The closest standard resistor is 1.89 kΩ so choose RUV1 = 1.89 kΩ.

Solve for RUVH given the hysteresis requirements:

Equation 56. TPS92690 Eq_slvsbk3_Ruvhclac.gif

The closest standard resistor is 14.3 kΩ so choose RUVH = 14.3 kΩ.

8.2.1.2.12 Output OVLO

Solve for ROV2:

Equation 57. TPS92690 Eq_slvsbk3_Rov2clac.gif

The closest standard resistor is 249 kΩ; therefore, choose RUV2 = 249 kΩ. Solve for ROV1:

Equation 58. TPS92690 Eq_slvsbk3_Rov1calc.gif

Choose the nearest standard resistor value of ROV1 = 8.06 kΩ.

8.2.1.3 Application Curve

TPS92690 g010.png Figure 23. Efficiency vs Input Voltage

8.2.2 Simplified Application

TPS92690 simp_app_slvsbk3.gif Figure 24. Simplified Application Schematic

8.2.2.1 Design Requirements

Number of series LEDs: N

Single LED forward voltage: VLED

Single LED dynamic resistance: rLED

Nominal input voltage: VIN

Input voltage range: VIN-MAX, VIN-MIN

Switching frequency: fSW

Current sense voltage: VCS

Average LED current: ILED

Inductor current ripple: ΔiL-PP

LED current ripple: ΔiLED-PP

Peak current limit: ILIM

Input voltage ripple: ΔvIN-PP

Output OVLO characteristics: VTURN-OFF, VHYSO

Input UVLO characteristics: VTURN-ON, VHYS

Total start-up time: tTSU

8.2.2.2 Detailed Design Procedure

8.2.2.2.1 Operating Point

Given the number of series LEDs (N), the forward voltage (VLED) and dynamic resistance (rLED) for a single LED, solve for the nominal output voltage (VO) and the nominal LED string dynamic resistance (rD):

Equation 59. TPS92690 30085786.gif
Equation 60. TPS92690 30085769.gif

Solve for the ideal nominal duty cycle (D):

Boost

Equation 61. TPS92690 30085714.gif

Buck-Boost

Equation 62. TPS92690 30085715.gif

Using the same equations, find the minimum duty cycle (DMIN) using maximum input voltage (VIN-MAX) and the maximum duty cycle (DMAX) using the minimum input voltage (VIN-MIN). Also, remember that D' = 1 - D.

8.2.2.2.2 Switching Frequency

Set the switching frequency (fSW) by solving for RT:

Equation 63. TPS92690 Eq_slvsbk3_RT.gif

8.2.2.2.3 Average LED Current

For all topologies, set the average LED current (ILED) knowing the desired current sense voltage (VCS) and solving for RCS:

Equation 64. TPS92690 Eq_slvsbk3_Vcs.gif
Equation 65. TPS92690 Eq_slvsbk3_Rcs.gif

If the calculated RCS is too far from a desired standard value, then VCS will have to be adjusted to obtain a standard value.

Setup the IADJ voltage by assuming RADJ2 = 100 kΩ and solving for RADJ1:

Equation 66. TPS92690 Eq_slvsbk3_Radj.gif

If the calculated RADJ1 is too far from a desired standard value, then RADJ2 can be adjusted to obtain a standard value.

8.2.2.2.4 Inductor Ripple Current

Find the minimum inductor value and calculate the nominal inductor ripple current (ΔiL-PP) by solving for the appropriate inductor (L1):

8.2.2.2.4.1 Minimum Inductor Value

Equation 67. TPS92690 Eq_slvsbk3_Lmin.gif

8.2.2.2.4.2 Inductor Ripple Current

Equation 68. TPS92690 eq_delta_vsbk3.gif

If the inductor ripple current is too high given the chosen value increase L1 to get the required inductor current ripple. For buck-boost applications replace VO with VIN + VO when solving for L1.

The minimum allowable inductor RMS current rating (IL-RMS) can be calculated as:

8.2.2.2.4.3 RMS Inductor Current

Equation 69. TPS92690 300857a9.gif

8.2.2.2.5 LED Ripple Current

Set the nominal LED ripple current (ΔiLED-PP), by solving for the output capacitance (CO):

8.2.2.2.5.1 Output Capacitor

Equation 70. TPS92690 30085719.gif

To set the worst case LED ripple current, use DMAX when solving for CO.

The minimum allowable RMS output capacitor current rating (ICO-RMS) can be approximated:

8.2.2.2.5.2 Output Capacitor RMS Current

Equation 71. TPS92690 30085721.gif

8.2.2.2.6 Peak Current Limit

Set the peak current limit (ILIM) by setting the ILIM pin voltage and solving for the transistor path sense resistor (RLIM):

Equation 72. TPS92690 Eq_slvsbk3_Vilim.gif
Equation 73. TPS92690 Eq_slvsbk3_Rlim.gif

8.2.2.2.7 Loop Compensation

Using a simple first order peak current mode control model, neglecting any output capacitor ESR dynamics, the necessary loop compensation can be determined.

A maximum bandwidth (fC) of 10 kHz is recommended and the COMP pin capacitor can be calculated using:

8.2.2.2.7.1 Compensation Capacitor

Equation 74. TPS92690 Eq_slvsbk3_Ccmp.gif

Check the location of the right-half plane zero and the output pole and make sure the crossover frequency is at least a decade below the lowest of the two using the following equations:

8.2.2.2.7.2 RHP Zero

Equation 75. TPS92690 eq_frhpz_vsbk3.gif

8.2.2.2.7.3 Output Capacitor Pole

Equation 76. TPS92690 Eq_slvsbk3_fpco.gif

If the input voltage range is wide use the maximum duty cycle (DMAX) corresponding to the minimum input voltage to calculate the RHP zero. In general smaller CCMP values will provide greater bandwidth but the bandwidth may be limited by the location of the RHP zero or output pole. For PWM dimming applications the largest capacitor value that will fit the applications requirements is suggested.

8.2.2.2.8 Input Capacitance

Set the nominal input voltage ripple (ΔvIN-PP) by solving for the required capacitance (CIN):

Boost

Equation 77. TPS92690 30085741.gif

Buck-Boost

Equation 78. TPS92690 30085742.gif

Use DMAX to set the worst case input voltage ripple.

The minimum allowable RMS input current rating (ICIN-RMS) can be approximated:

Boost

Equation 79. TPS92690 30085744.gif

Buck-Boost

Equation 80. TPS92690 30085745.gif

8.2.2.2.9 NFET

The NFET voltage rating should be at least 15% higher than the maximum NFET drain-to-source voltage (VT-MAX):

Boost

Equation 81. TPS92690 30085753.gif

Buck-Bosst

Equation 82. TPS92690 30085754.gif

The current rating should be at least 10% higher than the maximum average NFET current (IT-MAX):

8.2.2.2.9.1 Maximum Average NFET Current

Equation 83. TPS92690 30085763.gif

Approximate the nominal RMS transistor current (IT-RMS) :

8.2.2.2.9.2 RMS Transistor Current

Equation 84. TPS92690 30085765.gif

Given an NFET with on-resistance (RDS-ON), solve for the nominal power dissipation (PT):

Equation 85. TPS92690 30085767.gif

8.2.2.2.10 Diode

The Schottky diode voltage rating should be at least 15% higher than the maximum blocking voltage (VRD-MAX):

Boost

Equation 86. TPS92690 30085770.gif

Buck-Boost

Equation 87. TPS92690 30085771.gif

The current rating should be at least 10% higher than the maximum average diode current (ID-MAX):

8.2.2.2.10.1 Maximum Average Diode Current

Equation 88. TPS92690 30085773.gif

Replace DMAX with D in the ID-MAX equation to solve for the average diode current (ID). Given a diode with forward voltage (VFD), solve for the nominal power dissipation (PD):

Equation 89. TPS92690 30085781.gif

8.2.2.2.11 Output OVLO

The output OVLO is programmed with the turn-off threshold voltage (VTURN-OFF) and the desired hysteresis (VHYSO). To set VHYSO, solve for ROV2:

Equation 90. TPS92690 30085784.gif

To set VTURN-OFF, solve for ROV1:

Equation 91. TPS92690 30085785.gif

8.2.2.2.12 Input UVLO

For all topologies, input UVLO is programmed with the turn-on threshold voltage (VTURN-ON) and the desired hysteresis (VHYS).

Method 1: If no PWM dimming is required, a two resistor network can be used. To set VHYS, solve for RUV2:

Equation 92. TPS92690 30085782.gif

To set VTURN-ON, solve for RUV1:

Equation 93. TPS92690 30085783.gif

Method 2: If PWM dimming is required, a three resistor network is suggested. To set VTURN-ON, assume RUV2 = 10 kΩ and solve for RUV1 as in Method 1. To set VHYS, solve for RUVH:

Equation 94. TPS92690 300857c3.gif

8.2.2.2.13 Soft-Start

For all topologies, if soft-start is desired, find the start-up time without CSS (tSU):

Equation 95. TPS92690 30085780.gif

Then, if the desired total start-up time (tTSU) is larger than tSU, solve for the base start-up time (tSU-SS-BASE), assuming that a CSS greater than 40% of CCMP will be used:

Equation 96. TPS92690 300857j2.gif

Then solve for CSS:

Equation 97. TPS92690 300857j3.gif

8.2.2.2.14 PWM Dimming Method

PWM dimming can be performed several ways:

Method 1: Connect the dimming MosFET (Q3) with the drain to the nDIM pin and the source to GND. Apply an external PWM signal to the gate of QDIM. A pull down resistor may be necessary to properly turn off Q3.

Method 2: Connect the anode of a Schottky diode to the nDIM pin. Apply an external inverted PWM signal to the cathode of the same diode.

8.2.2.2.15 Analog Dimming Method

Analog dimming can be performed several ways:

Method 1: Place a potentiometer in place of RIADJ1.

Method 2: Connect a controlled voltage source to the IADJ pin to control the current sense voltage (VCS).