EMI occurs when electric or magnetic fields couple and interfere between two or more electronic devices or systems. In an electronic system, voltage ripple can result in conducted noise propagating from one circuit to another, especially when there are shared connections such as power-supply rails.

In a simple example, imagine hearing audible noise in a radio system that goes away when removing or replacing a faulty device. That device could have been generating ripple, causing interference within the audible range and coupling to the audio output.

In a broader sense, EMI is not limited to audible noise and can interfere with power, system inputs, processing and system outputs. International EMI standards such as Comité International Spécial des Perturbations Radioélectriques (CISPR) 25 specify EMI amplitude limits for different frequencies [1]. In Figure 2-1, conducted noise amplitude is represented on the Y axis in decibel microvolts; frequency is represented on the X axis in megahertz. The graph plots CISPR 25 noise-limit lines with peak-level detector limits in red and average levels in blue. Noise detected using a specific test setup and equipment specified by the CISPR standard must remain below these limit lines.

Figure 2-1 Conducted EMI plot with CISPR 25 limits.

In battery-powered systems (see Figure 3-1), a switched-mode power supply like a buck converter commutates a switch node between a low voltage (GND) and an input voltage. Filtering the switch-node voltage generates an average DC output voltage, which is between the input voltage and GND in the case of a buck converter. The switching causes input ripple at the fundamental switching frequency, and the square edges result in higher-frequency harmonics. Sharper edges with faster slew rates generate higher-frequency harmonic energy. The trapezoidal input current ripple is discontinuous and exhibits the same high-frequency components in the current ripple.

Parasitic capacitances and inductances are nonideal properties of components in the circuit. These parasitics interact negatively with the voltage and current and generate very-high-frequency spikes and ringing noise. On the other hand, jitter and dithering (spread spectrum) can create low-frequency oscillation and noise. Figure 3-2 shows common frequency ranges where noise occurs from these sources.

Figure 3-1 Simplified buck controller schematic and operating waveforms.

Figure 3-2 EMI frequency ranges for spread spectrum and jitter, switching frequency and harmonics, and switch ringing.

This document discusses advanced power-converter features that improve upon existing methods to further reduce EMI. The five features entail the use of:

• An active EMI filter (AEF) to improve passive EMI filter performance.
• Dual-random spread spectrum (DRSS) to improve triangular and random spread spectrum.
• True slew-rate control to improve the series boot resistor technique.
• HotRod™ package technology to improve standard wire-bond connections.
• Integrated capacitors to improve external decoupling capacitor performance.

Figure 4-1 illustrates how to mitigate and filter EMI with passive components.

Figure 4-1 Typical input filter components.

A high-frequency ceramic input capacitor helps supply the power metal-oxide semiconductor field-effect transistors (MOSFETs) with high-frequency energy to improve the switch’s slew-rate and edge characteristics, thus reducing switch ringing and high-frequency noise. Bulk capacitance provides low-frequency damping and prevents resonance between the filter components. A capacitor-inductor-capacitor (CLC) EMI filter, also known as a π filter, filters differential-mode noise from 10 MHz to 100 MHz, depending on the components selected.

Other components include a ferrite bead that provides high differential-mode impedance at very high frequencies. The ferrite bead impedance is typically specified at 100 MHz. A common-mode choke filters common-mode noise that usually occurs at high frequencies (10 MHz to 300 MHz).

Conventional CLC EMI filters use a capacitor and inductor arranged as a low-pass filter to attenuate noise from a noisy node to a quiet node. You can design the CLC components based on the required attenuation. Many tools are available to aid in designing filters, including application notes [2] and spreadsheet tools [3], as shown in Figure 4-2.

Figure 4-2 Input filter calculator Excel tool for designing a CLC EMI filter.

Alternatively, you could use Equation 1 to measure or estimate the required attenuation:

where f_SW is the switching frequency, C_IN is the input capacitance, D is the duty cycle, I_DC is the converter-inductor DC current and V_MAX is the decibel microvolt (dBµV) limit at the switching frequency.

Looking again at Figure 4-2 and considering the filter inductor (L_IN) and filter capacitor (C_F), the CLC π filter provides 40 dB of attenuation per decade at frequencies above the cutoff frequency. Figure 3-1 takes the attenuation required at the switching frequency to design the π-filter cutoff frequency:

You can then use Equation 3 to select filter components based on the required cutoff frequency:

After selecting a typical L_IN range between 0.1 µH and 10 µH, you can calculate the required C_F value. Increasing the capacitance value – as opposed to using larger inductance values, since a larger inductance tends to increase the overall filter size – lowers the cutoff frequency. Plus, higher-value inductors tend to have decreased current ratings for a given size. The most space-efficient designs use more parallel capacitors rather than a larger inductor value.

Increasing C_F capacitance with multiple parallel capacitors is usually easier and more space-effective than increasing L_IN to adjust the filter cutoff frequency. Furthermore, a higher inductance will have a decreased current rating for given size.

C_F connects to the main power rail and must be rated for the maximum DC voltage. Designs requiring a large C_F and a higher DC voltage will necessitate the use of a larger and more expensive capacitor. Similarly, L_IN must be rated for the maximum input current (I_IN) at minimum input voltages. In higher-power systems, the inductor size and cost will further increase given the larger inductance and DC current requirements. Transients such as load dump and cold crank exacerbate the V_IN and I_IN requirements.

A second-order effect is that the self-resonant frequency (SRF) decreases as the size of C_F increases. At the SRF, the impedance of the equivalent series inductance (ESL) equals that of the capacitance and marks the lowest impedance point over the frequency range. A higher ESL deteriorates the filter’s performance and its ability to attenuate noise at high frequencies.

Figure 5-1 shows two capacitors where the larger capacitor (in blue) is less effective at higher frequencies compared to the smaller capacitor (in black). The figure shows both the impedance and effective capacitance. At frequencies above the SRF, the capacitance rolls off and the parasitic inductance dominates the capacitor impedance. A similar SRF effect can limit L_IN in filter performance, where parasitic capacitance dominates the inductor and deteriorates performance at high frequencies.

Figure 5-1 Impedance of two different package and value capacitors: the larger capacitor (blue) is in a larger package and has a lower SRF, which results in a larger impedance at high frequencies compared to the smaller-package capacitor (black).

An AEF uses an amplifier circuit to sense noise on the input rail and inject an out-of-phase signal to cancel the noise being sensed. As shown in Figure 6-1, an operational-amplifier (op-amp) network acts effectively to replace a passive capacitor (C_F) with an active capacitor. The op-amp network requires feedback and compensation components, but these components are much smaller in size and cost than a large passive C_F. Integrating the op amp into a DC/DC controller package, as in the LM25149-Q1, results in a smaller overall filter size when compared to a fully passive EMI filter.

Figure 6-1 A passive EMI filter (a); an AEF-equivalent model (b); an AEF implementation (c).

Figure 6-2 shows a more detailed view of an AEF circuit. An AEF with voltage sensing and current injection (VSCI) uses capacitive sensing and injection; therefore, it does not source or sink any DC current. The AEF cancels the AC ripple current of the input filter inductor (as illustrated in Figure 6-2, with triangular waveforms for simplicity). Consequently, the DC input current does not determine the size of the AEF; instead, it is limited by the op amp’s source-and-sink current capability, as it cancels the AC ripple current in the filter inductor. As a result, an AEF can work with high levels of DC power, with appropriate sizes of L_IN and C_IN to keep the input ripple and noise within the range of the op amp’s source-and-sink current capability.

Figure 6-2 An AEF implementation in which capacitive sensing and injection ensure that the AEF does not source or sink DC current; as a result, the AEF size is independent of DC power, and only limited to canceling the input inductor ripple current.

The op amp forms a band-stop filter. The term G_Op expresses the gain of the op amp, approximated by the active EMI sensing and compensation capacitors C_SEN and C_AEFC in Equation 4:

Equation 5 has the same format as Equation 3; however, C_F is replaced with an injection capacitor (C_INJ) and G_Op. Using an AEF, you can design for a similar filter cutoff frequency or performance using much lower component values for L_IN and C_INJ.

For example, with G_Op equal to 100, an AEF can reduce the inductance of L_IN and the capacitance of C_INJ by a factor of 10 for each component. Furthermore, a smaller L_IN and C_INJ will have lower parasitics and much higher SRFs for better high-frequency performance.

Figure 6-3 and Table 6-1 compare passive and active EMI filters. The passive filter layout uses an inductor that has a significantly larger volume. The AEF layout uses smaller and cheaper components while achieving improved filter performance. The op amp is integrated in the controller integrated circuit (IC), not shown in Figure 6-3.

Figure 6-3 A passive EMI filter with two 1210 capacitors (a); an AEF with significantly smaller and cheaper components (b).

Figure 6-4 shows the EMI performance of the regulator – without a filter, with a passive EMI filter, and with an AEF. The AEF uses smaller components for a 50% area reduction, with improved EMI attenuation.

Figure 6-4 An EMI scan with no filter (a); an EMI scan with a passive EMI filter (b); an EMI scan with an AEF (c).

It is important to carefully select active EMI components, as some components can cause resonances if you do not meet certain frequency requirements. A sensing or injection component SRF that is less than the active EMI system crossover frequency can create unintended resonance, as shown in Figure 6-5. To avoid resonances, it is recommended to have an active EMI network crossover of around 15 MHz set by input compensation resistor and capacitance R_INC and C_INC, and using components with an SRF of at least 15 to 20 MHz.

Figure 6-5 An EMI scan with no filter (a); an EMI scan with an AEF using a low-quality injection capacitor with a component SRF of 5 MHz (b).

Component layout is an important consideration for AEFs. Keep loop areas formed by circuits small and away from noise sources, and connected directly to a ground plane where large currents are not flowing. At the op-amp IC, place the VCC decoupling capacitor very close to the VCC and power GND pins. Ensure that the AEF amplifier is grounded directly to the ground plane where high currents are not flowing. Keeping the loop area small, route the SEN and INJ traces tightly together from the amplifier.

As shown in Figure 6-6, the SEN and INJ traces should connect directly at compensation components R_AEFC, C_AEFC and R_AEFDC in the blue square. Place the sensing C_SEN close to the compensation and the injection-damping network R_DAMP, C_DAMP and C_INJ on the opposite side of sensing, as shown in the red and yellow squares. Finally, be sure that the input compensation C_INC and RINC connect to quiet GND, as shown in the purple squares.

Figure 6-6 Active EMI schematic (a); suggested layout (b).

Spread spectrum is a control method that dithers the switching frequency to spread the peak levels of EMI. Spread spectrum is most effective at reducing EMI peak levels at the switching frequency and harmonics, resulting in as much as 10 dBµV of reduction.

There are different methods of dithering the switching frequency for different frequency bands. The modulation frequency of dithering usually adds low-frequency noise in the EMI signature. Two traditional methods of modulating the switching frequency are triangular modulation and pseudo-random spread spectrum.

Figure 7-1 shows the modulation waveforms, switching timings and expected low-frequency EMI of both the triangular and pseudo-random spread-spectrum methods. Triangular spread spectrum dithers the switching frequency in a triangular pattern and spreads the fundamental frequency evenly because of the symmetric distribution. Pseudo-random dithers the switching frequency randomly after every switching cycle, which may not weigh each frequency evenly.

Figure 7-1 Illustration of triangular and pseudo-random spread-spectrum modulation waveform, switching waveform, and low-frequency EMI scan.

Triangular spread spectrum performs better at low frequencies because of the even distribution. Pseudo-random spread spectrum performs better in high-frequency EMI measurements, which are sampled in a span of time that may be shorter than the triangular modulation time. As a result, the full triangular modulation index will not modulate the sampled distribution of frequencies, resulting in more high-frequency EMI. See Figure 7-2.

Figure 7-2 Illustration of triangular and pseudo-random spread-spectrum modulation and the resulting high-frequency EMI scans. Because the resolution bandwidth is finer at high frequencies, triangular modulation does not spread the frequency as much as pseudo random within the smaller sample time, and has a worse high-frequency EMI signature. Note the margin between the scans and limit lines.

Triangular spread spectrum can introduce audible noise at the modulation frequency and its harmonics. Pseudo-random spread spectrum can also generate noise, but instead of a distinct tone, it is a distributed noise tone similar to white noise. Figure 7-3 illustrates the tone.

Figure 7-3 Illustration of triangular and pseudo-random spread-spectrum modulation and low-frequency EMI scans. Triangular spread spectrum will exhibit a tone at the modulation frequency.

DRSS combines triangular spread spectrum with random modulation of the triangular modulation frequency, with pseudo random added on top [4]. Figure 8-1 shows an example of a DRSS modulation profile. The combination of triangular and pseudo random takes the best from both schemes, while the additional modulation of the triangular modulation frequency spreads the audible tone.

Figure 8-1 A DRSS modulation waveform is a combination of triangular and pseudo random, while the triangular modulation frequency is randomly modulated as well.

DRSS will still have audible noise spread among the tone, thus necessitating the need for an active ripple cancellation scheme. The modulation of the switching frequency changes the converter’s ripple current because of how quickly the switching period can change compared to the time it takes for the control loop to adjust to the switching frequency change. The changing ripple current creates an average output-voltage ripple that follows the frequency modulation. The active ripple cancellation will scale the peak current command inverse to the switching frequency timing to reject this ripple.

Figure 8-2 shows both the modulation and output voltage ripple with active ripple cancellation on and off. Without ripple cancellation, the output voltage follows the modulation frequency.

Figure 8-2 Effect of spread spectrum on the output ripple: without ripple cancellation, the output ripple will follow the spread-spectrum modulation shown on the left; ripple cancellation shown on the right cancels this unwanted effect.

Slew-rate control is the ability to slow down the rising switch-node voltage and current in a buck regulator to reduce high-frequency emissions. At a high-enough frequency, the switch-node square wave will begin to look like a trapezoid wave with harmonics rolling off at 40 dB/dec, as shown in Figure 9-1. The ability to slow down the slew rate by controlling the drive strength of the high-side field-effect transistor’s (FET) driver enables the harmonics to roll off at lower frequencies, thereby reducing the overall noise amplitude measurement and providing more margin below violation limits in industry EMI standards.

Figure 9-1 EMI emission of a square wave.

One conventional way to control the slew rate in a buck regulator is to place a resistor in series with the boot capacitor, as shown in Figure 9-2. An unintended consequence of adding RBOOT is that the bootstrap voltage of the high-side driver may drop below its undervoltage lockout as peak gate-drive currents flow through it, affecting normal operation of the buck converter.

Figure 9-2 Conventional slew-rate control method.

Implemented in buck converters such as the LM61460-Q1 and LM61495-Q1, an improved slew-rate control method uses a dedicated pin (RBOOT). Figure 9-3 shows a true slew-rate control feature with a dedicated boot resistor that controls the drive strength of the high-side FET’s driver. A higher RBOOT resistance yields slower switch-node rise times. Accurately controlling the rise time of the switch-node voltage makes it possible to precisely control the switch-node harmonics’ rolloff frequency, which effectively improves the noise amplitude measurement. In some applications, true slew-rate control can eliminate the need for shielding and common-mode chokes, which further reduces total solution size.

Figure 9-3 The true slew-rate control method.

As tested, increasing the RBOOT value from 0 Ω to 300 Ω results in an approximate 0.5% drop in efficiency (as shown in Figure 9-4) but yields a solution that is CISPR 25 Class 5-compliant (as shown in Figure 9-5).

Figure 9-4 Efficiency comparisons with different RBOOT resistors.

Figure 9-5 CISPR 25 EMI comparisons with different RBOOT resistors.

TI’s buck-regulator portfolio provides a multitude of different package technologies that can help meet design expectations. The two most common package technologies available are standard wire-bond quad flat no-lead (QFN) and flip-chip-on-leadframe (the HotRod package). Packages optimized for EMI have reduced power-loop parasitic inductance, which reduces switch-node ringing.

As shown in Figure 10-1, QFN packages connect the die to the leadframe through wire bonds. A design drawback of the QFN package device is the additional parasitic inductance resulting from the wire bond connecting the copper bump and silicon die to the leadframe. Figure 10-2 shows the internal parasitic inductance (L_para3 and L_para4) in the simplified buck schematic. The parasitic inductance resonates with the parasitic capacitance of the switch node at every switch edge, causing undamped switch ringing and thus the potential for EMI.

Figure 10-1 A standard QFN package construction.

Figure 10-2 Typical buck converter input parasitic elements.

The HotRod package, as shown in Figure 10-3, provides a lower-noise solution by eliminating the use of bond wires. HotRod packages significantly minimize parasitic inductance, since the copper bumps are in direct contact with the leadframe. Step-down converters such as the LM61460-Q1, LMQ61460-Q1 and LM61495-Q1 are available in HotRod packaging to help meet system EMI requirements.

Figure 10-3 HotRod package construction.

Another benefit of HotRod package technology is the optimization of the input path pinouts. The input-high transient current (di/dt) loop is critical in a step-down converter; Figure 11-1 illustrates the importance of minimizing the area of this loop. HotRod package technology enables you to easily place and route the input decoupling capacitors close to the input and ground pins. The reduction in switching power-loop parasitic inductances and minimal input trace routing contribute to a lower EMI signature. Input pinout optimization also reduces switch-node ringing, output voltage noise and EMI.

Figure 11-1 Input capacitor placement near the converter IC (a) results in 7 dBµV less noise than placement farther away (b).

Current flowing through a copper trace generates a magnetic field, which results in an overall increase in EMI noise measurements. The HotRod package pinout is designed to have the input current loop split onto either side of the device, as shown in Figure 11-2. This symmetrical input and ground pinout consideration in HotRod devices creates an equal and opposite magnetic field, which provides a self-containing effect on the magnetic field and further reduces EMI [5].

Figure 11-2 Symmetrical HotRod package pinout.

The input current loop is a high di/dt loop that affects EMI at higher frequency ranges. Devices that integrate a high-frequency input decoupling capacitor effectively reduce the high di/dt inductive loop area and help further reduce EMI.

Figure 12-1 provides an example of a device with integrated capacitors. These integrated capacitors are soldered directly onto the internal leadframe of the device, which minimizes the parasitic inductance on the input loop.

Figure 12-1 Integrated capacitors in the LMQ61460-Q1.

The lab EMI measurements shown in Figure 12-2 demonstrate that, without an input ferrite bead, a device with an integrated capacitor provides approximately 8 dBµV of margin compared to a device without an integrated capacitor. When comparing a device with an integrated capacitor to a device without an integrated capacitor – but with an input ferrite bead – the device with the integrated capacitor provides an approximate 2- to 3-dBµV improvement.

Figure 12-2 EMI comparisons between the LM61460-Q1 and LMQ61460-Q1 automotive buck converters.

The LMQ61460-Q1 and LM62440-Q1 step-down converters use this device package construction method. Integrating the input decoupling capacitors inside the device provides a solution that is resistant to EMI and easy to lay out.

As shown in Figure 12-3, a device with integrated input decoupling capacitors lowers EMI and offers more margin without the need for slew-rate control compared to a device using only slew-rate control and no integrated capacitors. By combining the HotRod package with integrated capacitors, the LMQ61460-Q1 aims to attenuate high-frequency noise at the input path. Any additional device features to help attenuate EMI noise such as true slew-rate control add to a more reliable EMI design.

Figure 12-3 EMI comparisons between the LM61460-Q1 with slew-rate control and the LMQ61460-Q1.

1. Hegarty, Tim. “An Engineer’s Guide to Low EMI in DC/DC Regulators.” Texas Instruments e-book, literature No. SLYY208, 2021.
2. Martin, Alan. 2013. “AN-2162 Simple Success With Conducted EMI From DC-DC Converters.” Texas Instruments application report, literature No. SNVA489C, April 2013.
3. Texas Instruments. n.d. Buck Regulator Input Filter Capacitor for Conducted EMI Compliance. Accessed Nov. 30, 2021.
4. Ramadass, Yogesh, Ambreesh Tripathi, and Paul Curtis. “Time-Saving and Cost-Effective Innovations for EMI Reduction in Power Supplies.” Texas Instruments white paper, literature No. SLYY200, April 2021.
5. Ramadass, Yogesh, and Ambreesh Tripathi. “Advanced EMI Mitigation Techniques for Automotive Converters.” Texas Instruments Analog Design Journal article, literature No. SLYT789, 1Q 2020