具有集成 PGA 和基准的 ADS1120 4 通道、2kSPS、低功耗、16 位 ADC

Table 1. Noise in μV_RMS (μV_PP)
at AVDD = 3.3 V, AVSS = 0 V, Normal Mode, and Internal Reference = 2.048 V

Table 2. ENOB from RMS Noise (Noise-free Bits from Peak-to-Peak Noise)
at AVDD = 3.3 V, AVSS = 0 V, Normal Mode, and Internal Reference = 2.048 V

Table 3. Noise in μV_RMS (μV_PP) with PGA Disabled
at AVDD = 3.3 V, AVSS = 0 V, Normal Mode, and Internal Reference = 2.048 V

Table 4. ENOB from RMS Noise (Noise-free Bits from Peak-to-Peak Noise) with PGA Disabled
at AVDD = 3.3 V, AVSS = 0 V, Normal Mode, and Internal Reference = 2.048 V

Table 5. Noise in μV_RMS (μV_PP)
at AVDD = 3.3 V, AVSS = 0 V, Turbo Mode, and Internal Reference = 2.048 V

Table 6. ENOB from RMS Noise (Noise-Free Bits from Peak-to-Peak Noise)
at AVDD = 3.3 V, AVSS = 0 V, Turbo Mode, and Internal Reference = 2.048 V

Table 7. Noise in μV_RMS (μV_PP) with PGA Disabled
at AVDD = 3.3 V, AVSS = 0 V, Turbo Mode, and Internal Reference = 2.048 V

Table 8. ENOB from RMS Noise (Noise-Free Bits from Peak-to-Peak Noise) with PGA Disabled
at AVDD = 3.3 V, AVSS = 0 V, Turbo Mode, and Internal Reference = 2.048 V

8.3.1 Multiplexer

The device contains a very flexible input multiplexer, as shown in Figure 38. Either four single-ended signals, two differential signals, or a combination of two single-ended signals and one differential signal can be measured. The multiplexer is configured by four bits (MUX[3:0]) in the configuration register. When single-ended signals are measured, the negative ADC input (AIN_N) is internally connected to AVSS by a switch within the multiplexer. For system-monitoring purposes, the analog supply (AVDD – AVSS) / 4 or the currently-selected external reference voltage (V_(REFPx) – V_(REFNx)) / 4 can be selected as inputs to the ADC. The multiplexer also offers the possibility to route any of the two programmable current sources to any analog input (AINx) or to any dedicated reference pin (REFP0, REFN0).

Figure 38. Analog Input Multiplexer

Electrostatic discharge (ESD) diodes to AVDD and AVSS protect the inputs. To prevent the ESD diodes from turning on, the absolute voltage on any input must stay within the range provided by Equation 4:

If the voltages on the input pins have any potential to violate these conditions, external Schottky clamp diodes or series resistors may be required to limit the input current to safe values (see the Absolute Maximum Ratings table). Overdriving an unused input on the device may affect conversions taking place on other input pins. If any overdrive on unused inputs is possible, TI recommends clamping the signal with external Schottky diodes.

8.3.2 Low-Noise PGA

The device features a low-noise, low-drift, high input impedance, programmable gain amplifier (PGA). The PGA can be set to gains of 1, 2, 4, 8, 16, 32, 64, or 128. Three bits (GAIN[2:0]) in the configuration register are used to configure the gain. A simplified diagram of the PGA is shown in Figure 39. The PGA consists of two chopper-stabilized amplifiers (A1 and A2) and a resistor feedback network that sets the PGA gain. The PGA input is equipped with an electromagnetic interference (EMI) filter.

Figure 39. Simplified PGA Diagram

V_IN denotes the differential input voltage V_IN = (V_(AINP) – V_(AINN)). The gain of the PGA can be calculated with Equation 5:

Gain is changed inside the device using a variable resistor, R_G. The differential full-scale input voltage range (FSR) of the PGA is defined by the gain setting and the reference voltage used, as shown in Equation 6:

Table 9 shows the corresponding full-scale ranges when using the internal 2.048-V reference.

8.3.2.1 PGA Common-Mode Voltage Requirements

To stay within the linear operating range of the PGA, the input signals must meet certain requirements that are discussed in this section.

The outputs of both amplifiers (A1 and A2) in Figure 39 can not swing closer to the supplies (AVSS and AVDD) than 200 mV. If the outputs OUT_P and OUT_N are driven to within 200 mV of the supply rails, the amplifiers saturate and consequently become nonlinear. To prevent this nonlinear operating condition the output voltages must meet Equation 7:

Equation 7. AVSS + 0.2 V ≤ V_(OUTN), V_(OUTP) ≤ AVDD – 0.2 V

Translating the requirements of Equation 7 into requirements referred to the PGA inputs (AIN_P and AIN_N) is beneficial because there is no direct access to the outputs of the PGA. The PGA employs a symmetrical design, therefore the common-mode voltage at the output of the PGA can be assumed to be the same as the common-mode voltage of the input signal, as shown in Figure 40.

Figure 40. PGA Common-Mode Voltage

The common-mode voltage is calculated using Equation 8:

Equation 8. V_CM = ½ (V_(AINP) + V_(AINN)) = ½ (V_(OUTP) + V_(OUTN))

The voltages at the PGA inputs (AIN_P and AIN_N) can be expressed as Equation 9 and Equation 10:

Equation 9. V_(AINP) = V_CM + ½ V_IN

Equation 10. V_(AINN) = V_CM – ½ V_IN

The output voltages (V_(OUTP) and V_(OUTN)) can then be calculated as Equation 11 and Equation 12:

Equation 11. V_(OUTP) = V_CM + ½ Gain · V_IN

Equation 12. V_(OUTN) = V_CM – ½ Gain · V_IN

The requirements for the output voltages of amplifiers A1 and A2 (Equation 7) can now be translated into requirements for the input common-mode voltage range using Equation 11 and Equation 12, which are given in Equation 13 and Equation 14:

Equation 13. V_{CM (MIN)} ≥ AVSS + 0.2 V + ½ Gain · V_{IN (MAX)}

Equation 14. V_{CM (MAX)} ≤ AVDD – 0.2 V – ½ Gain · V_{IN (MAX)}

In order to calculate the minimum and maximum common-mode voltage limits, the maximum differential input voltage (V_{IN (MAX)}) that occurs in the application must be used. V_{IN (MAX)} can be less than the maximum possible FS value.

In addition to Equation 13, the minimum V_CM must also meet Equation 15 because of the specific design implementation of the PGA.

Equation 15. V_{CM (MIN)} ≥ AVSS + ¼ (AVDD – AVSS)

Figure 41 and Figure 42 show a graphical representation of the common-mode voltage limits for AVDD = 3.3 V and AVSS = 0 V, with gain = 1 and gain = 16, respectively.

AVDD = 3.3 V

Figure 41. Common-Mode Voltage Limits (Gain = 1)

AVDD = 3.3 V

Figure 42. Common-Mode Voltage Limits (Gain = 16)

The following discussion explains how to apply Equation 13 through Equation 15 to a hypothetical application. The setup for this example is AVDD = 3.3 V, AVSS = 0 V, and gain = 16, using an external reference,
V_ref = 2.5 V. The maximum possible differential input voltage V_IN = (V_(AINP) – V_(AINN)) that can be applied is then limited to the full-scale range of FSR = ±2.5 V / 16 = ±0.156 V. Consequently, Equation 13 through Equation 15 yield an allowed V_CM range of 1.45 V ≤ V_CM ≤ 1.85 V.

If the sensor signal connected to the inputs in this hypothetical application does not make use of the entire full-scale range but is limited to V_{IN (MAX)} = ±0.1 V, for example, then this reduced input signal amplitude relaxes the V_CM restriction to 1.0 V ≤ V_CM ≤ 2.3 V.

In the case of a fully-differential sensor signal, each input (AIN_P, AIN_N) can swing up to ±50 mV around the common-mode voltage (V_(AINP) + V_(AINN)) / 2, which must remain between the limits of 1.0 V and 2.3 V. The output of a symmetrical wheatstone bridge is an example of a fully-differential signal. Figure 43 shows a situation where the common-mode voltage of the input signal is at the lowest limit. V_(OUTN) is exactly at 0.2 V in this case. Any further decrease in common-mode voltage (V_CM) or increase in differential input voltage (V_IN) drives V_(OUTN) below 0.2 V and saturates amplifier A2.

Figure 43. Example where V_CM is at Lowest Limit

In contrast, the signal of an RTD is of a pseudo-differential nature (if implemented as shown in the RTD Measurement section), where the negative input is held at a constant voltage other than 0 V and only the voltage on the positive input changes. When a pseudo-differential signal must be measured, the negative input in this example must be biased at a voltage between 0.95 V and 2.25 V. The positive input can then swing up to
V_{IN (MAX)} = 100 mV above the negative input. Note that in this case the common-mode voltage changes at the same time the voltage on the positive input changes. That is, while the input signal swings between 0 V ≤ V_IN ≤ V_{IN (MAX)}, the common-mode voltage swings between V_(AINN) ≤ V_CM ≤ V_(AINN) + ½ V_{IN (MAX)}. Satisfying the common-mode voltage requirements for the maximum input voltage V_{IN (MAX)} ensures the requirements are met throughout the entire signal range.

Figure 44 and Figure 45 show examples of both fully-differential and pseudo-differential signals, respectively.

Figure 44. Fully-Differential Input Signal

Figure 45. Pseudo-Differential Input Signal

NOTE

Remember, common-mode voltage requirements with PGA enabled (Equation 13 to Equation 15) are as follows:

• V_{CM (MIN)} ≥ AVSS + ¼ (AVDD – AVSS)
• V_{CM (MIN)} ≥ AVSS + 0.2 V + ½ Gain · V_{IN (MAX)}
• V_{CM (MAX)} ≤ AVDD – 0.2 V – ½ Gain · V_{IN (MAX)}

8.3.3 Modulator

A ΔΣ modulator is used in the ADS1120 to convert the analog input voltage into a pulse code modulated (PCM) data stream. The modulator runs at a modulator clock frequency of f_(MOD) = f_(CLK) / 16 in normal and duty-cycle mode and f_(MOD) = f_(CLK) / 8 in turbo mode, where f_(CLK) is either provided by the internal oscillator or the external clock source. Table 10 shows the modulator frequency for each operating mode using either the internal oscillator or an external clock of 4.096 MHz.

8.3.4 Digital Filter

The device uses a linear-phase finite impulse response (FIR) digital filter that performs both filtering and decimation of the digital data stream coming from the modulator. The digital filter is automatically adjusted for the different data rates and always settles within a single cycle. At data rates of 5 SPS and 20 SPS, the filter can be configured to reject 50-Hz or 60-Hz line frequencies or to simultaneously reject 50 Hz and 60 Hz. Two bits (50/60[1:0]) in the configuration register are used to configure the filter accordingly. The frequency responses of the digital filter are shown in Figure 46 to Figure 59 for different output data rates using the internal oscillator or an external 4.096-MHz clock.

The filter notches and output data rate scale proportionally with the clock frequency. For example, a notch that appears at 20 Hz when using a 4.096-MHz clock appears at 10 Hz if a 2.048-MHz clock is used. Note that the internal oscillator can vary over temperature as specified in the Electrical Characteristics table. The data rate or conversion time, respectively, and filter notches consequently vary by the same amount. Consider using an external precision clock source if a digital filter notch at a specific frequency with a tighter tolerance is required.

Simultaneous 50-Hz and 60-Hz Rejection, 50/60[1:0] = 01

Figure 46. Filter Response
(DR = 20 SPS)

Simultaneous 50-Hz and 60-Hz Rejection, 50/60[1:0] = 01

Figure 47. Detailed View of Filter Response
(DR = 20 SPS)

50-Hz Rejection Only, 50/60[1:0] = 10

Figure 48. Filter Response
(DR = 20 SPS)

60-Hz Rejection Only, 50/60[1:0] = 11

Figure 50. Filter Response
(DR = 20 SPS)

50/60[1:0] = 00

Figure 52. Filter Response
(DR = 20 SPS)

Figure 54. Filter Response
(DR = 90 SPS)

Figure 56. Filter Response
(DR = 330 SPS)

Figure 58. Filter Response
(DR = 1 kSPS)

50-Hz Rejection Only, 50/60[1:0] = 10

Figure 49. Detailed View of Filter Response
(DR = 20 SPS)

60-Hz Rejection Only, 50/60[1:0] = 11

Figure 51. Detailed View of Filter Response
(DR = 20 SPS)

Figure 53. Filter Response
(DR = 45 SPS)

Figure 55. Filter Response
(DR = 175 SPS)

Figure 57. Filter Response
(DR = 600 SPS)

Figure 59. Filter Response
(DR = 2 kSPS)

8.3.5 Output Data Rate

Table 11 shows the actual conversion times for each data rate setting. The values provided are in terms of t_(CLK) cycles using an external clock with a clock frequency of f_(CLK) = 4.096 MHz. The data rates scale proportionally in case an external clock with a frequency other than 4.096 MHz is used.

Continuous conversion mode data rates are timed from one DRDY falling edge to the next DRDY falling edge. The first conversion starts 210 · t_(CLK) (normal mode, duty-cycle mode) or 114 · t_(CLK) (turbo mode) after the last SCLK falling edge of the START/SYNC command.

Single-shot mode data rates are timed from the last SCLK falling edge of the START/SYNC command to the DRDY falling edge and rounded to the next t_(CLK). In case the internal oscillator is used, an additional oscillator wake-up time of up to 50 µs (normal mode, duty-cycle mode) or 25 µs (turbo mode) must be added in single-shot mode. The internal oscillator starts to power up at the first SCLK rising edge of the START/SYNC command. If an SCLK frequency higher than 160 kHz (normal mode, duty-cycle mode) or 320 kHz (turbo mode) is used, the oscillator may not be fully powered up at the end of the START/SYNC command. The ADC then waits until the internal oscillator is fully powered up before starting a conversion.

Single-shot conversion times in duty-cycle mode are the same as in normal mode. See the Duty-Cycle Mode section for more details on duty-cycle mode operation.

Note that even though the conversion time at the 20-SPS setting is not exactly 1 / 20 Hz = 50 ms, this discrepancy does not affect the 50-Hz or 60-Hz rejection. To achieve the 50-Hz and 60-Hz rejection specified in the Electrical Characteristics, the external clock frequency must be 4.096 MHz. When using the internal oscillator, the conversion time and filter notches vary by the amount specified in the Electrical Characteristics table for oscillator accuracy.

8.3.6 Voltage Reference

The device offers an integrated low-drift, 2.048-V reference. For applications that require a different reference voltage value or a ratiometric measurement approach, the device offers two differential reference input pairs (REFP0, REFN0 and REFP1, REFN1). In addition, the analog supply (AVDD – AVSS) can be used as a reference.

The reference source is selected by two bits (VREF[1:0]) in the configuration register. By default, the internal reference is selected. The internal voltage reference requires less than 25 µs to fully settle after power-up, when coming out of power-down mode, or when switching from an external reference source to the internal reference.

The differential reference inputs allow freedom in the reference common-mode voltage. REFP0 and REFN0 are dedicated reference inputs whereas REFP1 and REFN1 are shared with inputs AIN0 and AIN3, respectively. All reference inputs are internally buffered to increase input impedance. Therefore, additional reference buffers are usually not required when using an external reference. When used in ratiometric applications, the reference inputs do not load the external circuitry. Note that the analog supply current increases when using an external reference because the reference buffers are enabled.

In most cases the conversion result is directly proportional to the stability of the reference source. Any noise and drift of the voltage reference is reflected in the conversion result.

8.3.7 Clock Source

The device system clock can either be provided by the internal low-drift oscillator or by an external clock source on the CLK input. Connect the CLK pin to DGND before power-up or reset to activate the internal oscillator. Connecting an external clock to the CLK pin at any time deactivates the internal oscillator after two rising edges on the CLK pin are detected. The device then operates on the external clock. After the ADS1120 switches to the external clock, the device can only be switched back to the internal oscillator by cycling the power supplies or by sending a RESET command.

8.3.8 Excitation Current Sources

The device provides two matched programmable excitation current sources (IDACs) for RTD applications. The output current of the current sources can be programmed to 50 μA, 100 μA, 250 μA, 500 μA, 1000 μA, or 1500 μA using the respective bits (IDAC[2:0]) in the configuration register. Each current source can be connected to any of the analog inputs (AINx) as well as to any of the dedicated reference inputs (REFP0 and REFN0). Both current sources can also be connected to the same pin. Routing of the IDACs is configured by bits (I1MUX[2:0], I2MUX[2:0]) in the configuration register. Care must be taken not to exceed the compliance voltage of the IDACs. In other words, limit the voltage on the pin where the IDAC is routed to ≤ (AVDD – 0.9 V), otherwise the specified accuracy of the IDAC current is not met. For three-wire RTD applications, the matched current sources can be used to cancel errors caused by sensor lead resistance (see the 3-Wire RTD Measurement section for more details).

The IDACs require up to 200 µs to start up after the IDAC current is programmed to the respective value using bits IDAC[2:0]. If configuration registers 2 and 3 are not written during the same WREG command, TI recommends to first set the IDAC current to the respective value using bits IDAC[2:0] and thereafter select the routing for each IDAC (I1MUX[2:0], I2MUX[2:0]).

In single-shot mode, the IDACs remain active between any two conversions if the IDAC[2:0] bits are set to a value other than 000. However, the IDACs are powered down whenever the POWERDOWN command is issued.

Note that the analog supply current increases when enabling the IDACs (that is, when the IDAC[2:0] bits are set to a value other than 000). The IDAC circuit needs this bias current to operate even when the IDACs are not routed to any pin (I1MUX[2:0] = I2MUX[2:0] = 000). In addition, the selected output current is drawn from the analog supply when I1MUX[2:0] or I2MUX[2:0] are set to a value other than 000.

8.3.9 Low-Side Power Switch

A low-side power switch with low on-resistance connected between the analog input AIN3/REFN1 and AVSS is integrated in the device as well. This power switch can be used to reduce system power consumption in bridge sensor applications by powering down the bridge circuit between conversions. When the respective bit (PSW) in the configuration register is set, the switch automatically closes when the START/SYNC command is sent and opens when the POWERDOWN command is issued. Note that the switch stays closed between conversions in single-shot mode in case the PSW bit is set to 1. The switch can be opened at any time by setting the PSW bit to 0. By default, the switch is always open.

8.3.10 Sensor Detection

To help detect a possible sensor malfunction, the device provides internal 10-µA, burn-out current sources. When enabled by setting the respective bit (BCS) in the configuration register, one current source sources current to the positive analog input (AIN_P) currently selected while the other current source sinks current form the selected negative analog input (AIN_N).

In case of an open circuit in the sensor, these burn-out current sources pull the positive input towards AVDD and the negative input towards AVSS, resulting in a full-scale reading. A full-scale reading may also indicate that the sensor is overloaded or that the reference voltage is absent. A near-zero reading may indicate a shorted sensor. Note that the absolute value of the burn-out current sources typically varies by ±10% and the internal multiplexer adds a small series resistance. Therefore, distinguishing a shorted sensor condition from a normal reading can be difficult, especially if an RC filter is used at the inputs. In other words, even if the sensor is shorted, the voltage drop across the external filter resistance and the residual resistance of the multiplexer causes the output to read a value higher than zero.

Keep in mind that ADC readings of a functional sensor may be corrupted when the burn-out current sources are enabled. TI recommends disabling the burn-out current sources when preforming the precision measurement, and only enabling them to test for sensor fault conditions.

8.3.11 System Monitor

The device provides some means for monitoring the analog power supply and the external voltage reference. To select a monitoring voltage, the internal multiplexer (MUX[3:0]) must be configured accordingly in the configuration register. The device automatically bypasses the PGA and sets the gain to 1, irrespective of the configuration register settings while the monitoring feature is used. Note that the system monitor function only provides a coarse result and is not meant to be a precision measurement.

When measuring the analog power supply (MUX[3:0] = 1101), the resulting conversion is approximately (AVDD – AVSS) / 4. The device uses the internal 2.048-V reference for the measurement regardless of what reference source is selected in the configuration register (VREF[1:0]).

When monitoring one of the two possible external reference voltage sources (MUX[3:0] = 1100), the result is approximately (V_(REFPx) – V_(REFNx)) / 4. REFPx and REFNx denote the external reference input pair selected in the configuration register (VREF[1:0]). The device automatically uses the internal reference for the measurement.

8.3.12 Offset Calibration

The internal multiplexer offers the option to short both PGA inputs (AIN_P and AIN_N) to mid-supply (AVDD + AVSS) / 2. This option can be used to measure and calibrate the device offset voltage by storing the result of the shorted input voltage reading in a microcontroller and consequently subtracting the result from each following reading. TI recommends taking multiple readings with the inputs shorted and averaging the result to reduce the effect of noise.

8.3.13 Temperature Sensor

The ADS1120 offers an integrated precision temperature sensor. The temperature sensor mode is enabled by setting bit TS = 1 in the configuration register. When in temperature sensor mode, the settings of configuration register 0 have no effect and the device uses the internal reference for measurement, regardless of the selected voltage reference source. Temperature readings follow the same process as the analog inputs for starting and reading conversion results. Temperature data are represented as a 14-bit result that is left-justified within the 16-bit conversion result. Data are output starting with the most significant byte (MSB). When reading the two data bytes, the first 14 bits are used to indicate the temperature measurement result. One 14-bit LSB equals 0.03125°C. Negative numbers are represented in binary twos complement format, as shown in Table 12.

8.4.1 Power-Up and Reset

When the device powers up, a reset is performed. The reset process takes approximately 50 µs. After this power-up reset time, all internal circuitry (including the voltage reference) are stable and communication with the device is possible. As part of the reset process, the device sets all bits in the configuration registers to the respective default settings. By default, the device is set to single-shot mode. After power-up, the device performs a single conversion using the default register settings and then enters a low-power state. When the conversion is complete, the DRDY pin transitions from high to low. The high-to-low transition of the DRDY pin can be used to signal that the ADS1120 is operational and ready to use. The power-up behavior is intended to prevent systems with tight power-supply requirements from encountering a current surge during power-up.

8.4.2 Conversion Modes

The device can be operated in one of two conversion modes that can be selected by the CM bit in the configuration register. These conversion modes are single-shot and continuous conversion mode.

8.4.3 Operating Modes

In addition to the different conversion modes, the device can also be operated in different operating modes that can be selected to trade-off power consumption, noise performance, and output data rate. These modes are: normal mode, duty-cycle mode, turbo mode, and power-down mode.

8.5.1 Serial Interface

The SPI-compatible serial interface of the device is used to read conversion data, read and write the device configuration registers, and control device operation. Only SPI mode 1 (CPOL = 0, CPHA = 1) is supported. The interface consists of five control lines (CS, SCLK, DIN, DOUT/DRDY, and DRDY) but can be used with only four or even three control signals as well. The dedicated data-ready signal (DRDY) can be configured to be shared with DOUT/DRDY. If the serial bus is not shared with any other device, CS can be tied low permanently so that only signals SCLK, DIN, and DOUT/DRDY are required to communicate with the device.

8.5.2 Data Format

The device provides 16 bits of data in binary twos complement format. The size of one code (LSB) is calculated using Equation 16.

A positive full-scale input [V_IN ≥ (+FS – 1 LSB) = (V_ref / Gain – 1 LSB)] produces an output code of 7FFFh and a negative full-scale input (V_IN ≤ –FS = –V_ref / Gain) produces an output code of 8000h. The output clips at these codes for signals that exceed full-scale.

Table 13 summarizes the ideal output codes for different input signals.

Mapping of the analog input signal to the output codes is shown in Figure 60.

Figure 60. Code Transition Diagram

8.5.3 Commands

The device offers six different commands to control device operation, as shown in Table 14. Four commands are stand-alone instructions (RESET, START/SYNC, POWERDOWN, and RDATA). The commands to read (RREG) and write (WREG) configuration register data from and to the device require additional information as part of the instruction.

8.5.4 Reading Data

Output pins DRDY and DOUT/DRDY (if the DRDYM bit is set high in the configuration register) transition low when new data are ready for retrieval. The conversion data are written to an internal data buffer. Data can be read directly from this buffer on DOUT/DRDY when DRDY falls low without concern of data corruption. An RDATA command does not have to be sent. Data are shifted out on the SCLK rising edges, MSB first, and consist of two bytes of data.

Figure 61 to Figure 63 show the timing diagrams for reading conversion data in continuous conversion mode and single-shot mode when not using the RDATA command.

Figure 61. Continuous Conversion Mode (DRDYM = 0)

Figure 62. Continuous Conversion Mode (DRDYM = 1)

Figure 63. Single-Shot Mode (DRDYM = 0)

Data can also be read at any time without synchronizing to the DRDY signal using the RDATA command. When an RDATA command is issued, the conversion result currently stored in the data buffer is shifted out on DOUT/DRDY on the following SCLK rising edges. Data can be read continuously with the RDATA command as an alternative to monitoring DRDY or DOUT/DRDY. The DRDY pin can be polled after the LSB is clocked out to determine if a new conversion result was loaded. If a new conversion completes during the read operation but data from the previous conversion are read, then DRDY is low. Otherwise, if the most recent result is read, DRDY is high. Figure 64 and Figure 65 illustrate the behavior for both cases.

Figure 64. State of DRDY when a New Conversion Finishes During an RDATA Command

Figure 65. State of DRDY when the Most Recent Conversion Result is Read During an RDATA Command

8.5.5 Sending Commands

The device serial interface is capable of full-duplex operation while reading conversion data when not using the RDATA command. Full-duplex operation means commands are decoded at the same time that conversion data are read. Commands can be sent on any 8-bit data boundary during a data read operation. When a RREG or RDATA command is recognized, the current data read operation is aborted and the conversion data are corrupted, unless the command is sent while the last byte of the conversion result is retrieved. The device starts to output the requested data on DOUT/DRDY at the first SCLK rising edge after the command byte. To read data without interruption, keep DIN low while clocking out data.

A WREG command can be sent without corrupting an ongoing read operation. Figure 66 shows an example for sending a WREG command to write two configuration registers while reading conversion data in continuous conversion mode. After the command is clocked in (after the 32nd SCLK falling edge), the device resets the digital filter and starts converting with the new register settings. The WREG command can be sent on any of the 8-bit boundaries.

Figure 66. Example of Reading Data while Simultaneously Sending a WREG Command

Note that the serial interface does not decode commands while an RDATA or RREG command is executed. That is, all 16 bits of the conversion result must be read after the RDATA command is issued and all requested registers must be read after a RREG command is sent before a new command can be issued.

8.5.6 Interfacing with Multiple Devices

When connecting multiple ADS1120 devices to a single SPI bus, SCLK, DIN, and DOUT/DRDY can be safely shared by using a dedicated chip-select (CS) line for each SPI-enabled device. When CS transitions high for the respective device, DOUT/DRDY enters a 3-state mode. Therefore, DOUT/DRDY cannot be used to indicate when new data are available if CS is high, regardless of the DRDYM bit setting in the configuration register. Only the dedicated DRDY pin indicates that new data are available, because the DRDY pin is actively driven even when CS is high.

In some cases the DRDY pin cannot be interfaced to the microcontroller. This scenario can occur if there are insufficient GPIO channels available on the microcontroller or if the serial interface must be galvanically isolated and thus the amount of channels must be limited. Therefore, in order to evaluate when a new conversion of one of the devices is ready, the microcontroller can periodically drop CS to the respective device and poll the state of the DOUT/DRDY pin. When CS goes low, the DOUT/DRDY pin immediately drives either high or low, provided that the DRDYM bit is configured to 1. If the DOUT/DRDY line drives low, when CS is taken low, new data are currently available. If the DOUT/DRDY line drives high, no new data are available. This procedure requires that DOUT/DRDY is high after reading each conversion result and before taking CS high. To make sure DOUT/DRDY is taken high, send 16 additional SCLKs with DIN held low after each data read operation. DOUT/DRDY reads low during the first eight SCLKs after the conversion result is read, and reads high during the following eight SCLKs, as shown in Figure 67. Alternatively, valid data can be retrieved from the device at any time without concern of data corruption by using the RDATA command.

Figure 67. Example of Taking DOUT/DRDY High After Reading a Conversion Result

8.6.1 Configuration Registers

The device has four 8-bit configuration registers that are accessible through the serial interface using the RREG and WREG commands. The configuration registers control how the device operates and can be changed at any time without causing data corruption. After power-up or reset, all registers are set to the default values (which are all 0). All registers retain their values during power-down mode. Table 15 shows the register map of the configuration registers.

9.1.1 Serial Interface Connections

The principle serial interface connections for the ADS1120 are shown in Figure 72.

Figure 72. Serial Interface Connections

Most microcontroller SPI peripherals can operate with the ADS1120. The interface operates in SPI mode 1 where CPOL = 0 and CPHA = 1. In SPI mode 1, SCLK idles low and data are launched or changed only on SCLK rising edges; data are latched or read by the master and slave on SCLK falling edges. Details of the SPI communication protocol employed by the device can be found in the SPI Timing Requirements section.

TI recommends placing 47-Ω resistors in series with all digital input and output pins (CS, SCLK, DIN, DOUT/DRDY, and DRDY). This resistance smooths sharp transitions, suppresses overshoot, and offers some overvoltage protection. Care must be taken to meet all SPI timing requirements because the additional resistors interact with the bus capacitances present on the digital signal lines.

9.1.2 Analog Input Filtering

Analog input filtering serves two purposes: first, to limit the effect of aliasing during the sampling process and second, to reduce external noise from being a part of the measurement.

As with any sampled system, aliasing can occur if proper antialias filtering is not in place. Aliasing occurs when frequency components are present in the input signal that are higher than half the sampling frequency of the ADC (also known as the Nyquist frequency). These frequency components are folded back and show up in the actual frequency band of interest below half the sampling frequency. Note that inside a ΔΣ ADC, the input signal is sampled at the modulator frequency f_(MOD) and not at the output data rate. The filter response of the digital filter repeats at multiples of the sampling frequency (f_(MOD)), as shown in Figure 73. Signals or noise up to a frequency where the filter response repeats are attenuated to a certain amount by the digital filter depending on the filter architecture. Any frequency components present in the input signal around the modulator frequency or multiples thereof are not attenuated and alias back into the band of interest, unless attenuated by an external analog filter.

Figure 73. Effect of Aliasing

Many sensor signals are inherently bandlimited; for example, the output of a thermocouple has a limited rate of change. In this case the sensor signal does not alias back into the pass-band when using a ΔΣ ADC. However, any noise pick-up along the sensor wiring or the application circuitry can potentially alias into the pass-band. Power line-cycle frequency and harmonics are one common noise source. External noise can also be generated from electromagnetic interference (EMI) or radio frequency interference (RFI) sources, such as nearby motors and cellular phones. Another noise source typically exists on the printed circuit board (PCB) itself in the form of clocks and other digital signals. Analog input filtering helps remove unwanted signals from affecting the measurement result.

A first-order resistor-capacitor (RC) filter is (in most cases) sufficient to either totally eliminate aliasing, or to reduce the effect of aliasing to a level within the noise floor of the sensor. Ideally, any signal beyond f_(MOD) / 2 is attenuated to a level below the noise floor of the ADC. The digital filter of the ADS1120 attenuates signals to a certain degree, as illustrated in the filter response plots in the Digital Filter section. In addition, noise components are usually smaller in magnitude than the actual sensor signal. Therefore, using a first-order RC filter with a cutoff frequency set at the output data rate or 10x higher is generally a good starting point for a system design.

Internal to the device, prior to the PGA inputs, is an EMI filter as shown in Figure 39. The cutoff frequency of this filter is approximately 31.8 MHz, which helps reject high-frequency interferences.

9.1.3 External Reference and Ratiometric Measurements

The full-scale range of the ADS1120 is defined by the reference voltage and the PGA gain (FSR = ±V_ref / Gain). An external reference can be used instead of the integrated 2.048-V reference to adapt the FSR to the specific system needs. An external reference must be used if V_IN > 2.048 V. For example, an external 5-V reference and an AVDD = 5 V are required in order to measure a single-ended signal that can swing between 0 V and 5 V.

The reference inputs of the device also allow the implementation of ratiometric measurements. In a ratiometric measurement the same excitation source that is used to excite the sensor is also used to establish the reference for the ADC. As an example, a simple form of a ratiometric measurement uses the same current source to excite both the resistive sensor element (such as an RTD) and another resistive reference element that is in series with the element being measured. The voltage that develops across the reference element is used as the reference source for the ADC. Because current noise and drift are common to both the sensor measurement and the reference, these components cancel out in the ADC transfer function. The output code is only a ratio of the sensor element and the value of the reference resistor. The value of the excitation current source itself is not part of the ADC transfer function.

9.1.4 Establishing a Proper Common-Mode Input Voltage

The ADS1120 can be used to measure various types of input signal configurations: single-ended, pseudo-differential, and fully-differential signals (which can be either unipolar or bipolar). However, configuring the device properly for the respective signal type is important.

Signals where the negative analog input is fixed and referenced to analog ground (V_(AINN) = 0 V) are commonly called single-ended signals. The common-mode voltage of a single-ended signal consequently varies between 0 V and V_IN / 2. If the PGA is disabled and bypassed, the common-mode input voltage of the ADS1120 can be as low as 100 mV below AVSS and as large as 100 mV above AVDD. Therefore, the PGA_BYPASS bit must be set in order to measure single-ended signals when a unipolar analog supply is used (AVSS = 0 V). Gains of 1, 2, and 4 are still possible in this configuration. Measuring a 0-mA to 20-mA or 4-mA to 20-mA signal across a load resistor of 100 Ω referenced to GND is a typical example. The ADS1120 can directly measure the signal across the load resistor using a unipolar supply, the internal 2.048-V reference, and gain = 1 when the PGA is bypassed.

If gains larger than 4 are needed to measure a single-ended signal, the PGA must be enabled. In this case, a bipolar supply is required for the ADS1120 to meet the common-mode voltage requirement of the PGA.

Signals where the negative analog input (AIN_N) is fixed at a voltage other the 0 V are referred to as pseudo-differential signals. The common-mode voltage of a pseudo-differential signal varies between V_(AINN) and V_(AINN) + V_IN / 2.

Fully-differential signals in contrast are defined as signals having a constant common-mode voltage where the positive and negative analog inputs swing 180° out-of-phase but have the same amplitude.

The ADS1120 can measure pseudo-differential and fully-differential signals both with the PGA enabled or bypassed. However, the PGA must be enabled in order to use gains greater than 4. The common-mode voltage of the input signal must meet the input-common mode voltage restrictions of the PGA (as explained in the PGA Common-Mode Voltage Requirements section) when the PGA is enabled. Setting the common-mode voltage at or near (AVSS + AVDD) / 2 in most cases satisfies the PGA common-mode voltage requirements.

Signals where both the positive and negative inputs are always ≥ 0 V are called unipolar signals. These signals can in general be measured with the ADS1120 using a unipolar analog supply (AVSS = 0 V). As mentioned previously, the PGA must be bypassed in order to measure single-ended, unipolar signals when using a unipolar supply.

A signal is called bipolar when either the positive or negative input can swing below 0 V. A bipolar analog supply (such as AVDD = 2.5 V, AVSS = –2.5 V) is required in order to measure bipolar signals with the ADS1120. A typical application task is measuring a single-ended, bipolar ±10 V signal where AIN_N is fixed at 0 V while AIN_P swings between –10 V and 10 V. The ADS1120 cannot directly measure this signal because the 10 V exceeds the analog power-supply limits. However, one possible solution is to use a bipolar analog supply (AVDD = 2.5 V, AVSS = –2.5 V), gain = 1, and a resistor divider in front of the ADS1120. The resistor divider must divide the voltage down to ≤ ±2.048 V to be able to measure it using the internal 2.048-V reference.

9.1.5 Unused Inputs and Outputs

To minimize leakage currents on the analog inputs, leave unused analog and reference inputs floating, or connect the inputs to mid-supply or to AVDD. AIN3/REFN1 is an exception. Leave the AIN3/REFN1 pin floating when not used in order to avoid accidently shorting the pin to AVSS through the internal low-side switch. Connecting unused analog or reference inputs to AVSS is possible as well, but can yield higher leakage currents than the previously mentioned options.

Do not float unused digital inputs; excessive power-supply leakage current can result. Tie all unused digital inputs to the appropriate levels, DVDD or DGND, even when in power-down mode. If CS is not used, tie this pin to DGND. If the internal oscillator is used, tie the CLK pin to DGND. If the DRDY output is not used, leave the pin unconnected or tie the pin to DVDD using a weak pullup resistor.

9.1.6 Pseudo Code Example

The following list shows a pseudo code sequence with the required steps to set up the device and the microcontroller that interfaces to the ADC in order to take subsequent readings from the ADS1120 in continuous conversion mode. The dedicated DRDY pin is used to indicate availability of new conversion data. The default configuration register settings are changed to gain = 16, continuous conversion mode, and simultaneous 50-Hz and 60-Hz rejection.

TI recommends running an offset calibration before performing any measurements or when changing the gain of the PGA. The internal offset of the device can, for example, be measured by shorting the inputs to mid-supply (MUX[3:1] = 1110). The microcontroller then takes multiple readings from the device with the inputs shorted and stores the average value in the microcontroller memory. When measuring the sensor signal, the microcontroller then subtracts the stored offset value from each device reading to obtain an offset compensated result. Note that the offset can be either positive or negative in value.

9.2.1 K-Type Thermocouple Measurement (–200°C to +1250°C)

Figure 74 shows the basic connections of a thermocouple measurement system when using the internal high-precision temperature sensor for cold-junction compensation. Apart from the thermocouple itself, the only external circuitry required are two biasing resistors, a simple low-pass, antialiasing filter, and the power-supply decoupling capacitors.

Figure 74. Thermocouple Measurement

9.2.1.3 Application Curves

Figure 75 and Figure 76 show the measurement results. The measurements are taken at T_A = T_(CJ) = 25°C. A system offset calibration is performed at T_(TC) = 25°C, which translates to a V_(TC) = 0 V when T_(CJ) = 25°C. No gain calibration is implemented. The data in Figure 75 are taken using a precision voltage source as the input signal instead of a thermocouple. The respective temperature measurement error in Figure 76 is calculated from the data in Figure 75 using the NIST tables.

The design meets the required temperature measurement accuracy given in Table 21. Note that the measurement error shown in Figure 76 does not include the error of the thermocouple itself and the measurement error of the cold-junction temperature. Those two error sources are in general larger than 0.2°C and therefore, in many cases, dominate the overall system measurement accuracy.

Figure 75. Voltage Measurement Error vs V_(TC)

Figure 76. Temperature Measurement Error vs T_(TC)

9.2.2 3-Wire RTD Measurement (–200°C to +850°C)

The ADS1120 integrates all necessary features (such as dual-matched programmable current sources, buffered reference inputs, and a PGA) to ease the implementation of ratiometric 2-, 3-, and 4-wire RTD measurements. Figure 77 shows a typical implementation of a ratiometric 3-wire RTD measurement using the excitation current sources integrated in the device to excite the RTD as well as to implement automatic RTD lead-resistance compensation.

Figure 77. 3-Wire RTD Measurement

9.2.2.2 Detailed Design Procedure

The circuit in Figure 77 employs a ratiometric measurement approach. In other words, the sensor signal (that is, the voltage across the RTD in this case) and the reference voltage for the ADC are derived from the same excitation source. Therefore, errors resulting from temperature drift or noise of the excitation source cancel out because these errors are common to both the sensor signal and the reference.

In order to implement a ratiometric 3-wire RTD measurement using the device, IDAC1 is routed to one of the leads of the RTD and IDAC2 is routed to the second RTD lead. Both currents have the same value, which is programmable by the IDAC[2:0] bits in the configuration register. The design of the device ensures that both IDAC values are closely matched, even across temperature. The sum of both currents flows through a precision, low-drift reference resistor, R_REF. The voltage, V_ref, generated across the reference resistor (as shown in Equation 20) is used as the ADC reference voltage. Equation 20 reduces to Equation 21 because I_IDAC1 = I_IDAC2.

Equation 20. V_ref = (I_IDAC1 + I_IDAC2) · R_REF

Equation 21. V_ref = 2 · I_IDAC1 · R_REF

To simplify the following discussion, the individual lead resistance values of the RTD (R_LEADx) are set to zero. Only IDAC1 excites the RTD to produce a voltage (V_RTD) proportional to the temperature-dependable RTD value and the IDAC1 value, as shown in Equation 22.

Equation 22. V_RTD = R_{RTD (at temperature)} · I_IDAC1

The device internally amplifies the voltage across the RTD using the PGA and compares the resulting voltage against the reference voltage to produce a digital output code proportional to Equation 23 through Equation 25:

Equation 23. Code ∝ V_RTD · Gain / V_ref

Equation 24. Code ∝ (R_{RTD (at temperature)} · I_IDAC1 · Gain) / (2 · I_IDAC1 · R_REF)

Equation 25. Code ∝ (R_{RTD (at temperature)} · Gain) / (2 · R_REF)

As can be seen from Equation 25, the output code only depends on the value of the RTD, the PGA gain, and the reference resistor (R_REF), but not on the IDAC1 value. The absolute accuracy and temperature drift of the excitation current therefore does not matter. However, because the value of the reference resistor directly affects the measurement result, choosing a reference resistor with a very low temperature coefficient is important to limit errors introduced by the temperature drift of R_REF.

The second IDAC2 is used to compensate for errors introduced by the voltage drop across the lead resistance of the RTD. All three leads of a 3-wire RTD typically have the same length and, thus, the same lead resistance. Also, IDAC1 and IDAC2 have the same value. Taking the lead resistance into account, the differential voltage (V_IN) across the ADC inputs, AIN0 and AIN1, is calculated using Equation 26:

Equation 26. V_IN = I_IDAC1 · (R_RTD + R_LEAD1) – I_IDAC2 · R_LEAD2

When R_LEAD1 = R_LEAD2 and I_IDAC1 = I_IDAC2, Equation 26 reduces to Equation 27:

Equation 27. V_IN = I_IDAC1 · R_RTD

In other words, the measurement error resulting from the voltage drop across the RTD lead resistance is compensated, as long as the lead resistance values and the IDAC values are well matched.

A first-order differential and common-mode RC filter (R_F1, R_F2, C_DIF1, C_CM1, and C_CM2) is placed on the ADC inputs, as well as on the reference inputs (R_F3, R_F4, C_DIF2, C_CM3, and C_CM4). The same guidelines for designing the input filter apply as described in the Thermocouple Measurement section. For best performance, TI recommends matching the corner frequencies of the input and reference filter. More detailed information on matching the input and reference filter can be found in application report RTD Ratiometric Measurements and Filtering Using the ADS1148 and ADS1248 (SBAA201).

The reference resistor R_REF not only serves to generate the reference voltage for the device, but also sets the common-mode voltage of the RTD to within the specified common-mode voltage range of the PGA.

When designing the circuit, care must also be taken to meet the compliance voltage requirement of the IDACs. The IDACs require that the maximum voltage drop developed across the current path to AVSS be equal or less than AVDD – 0.9 V in order to operate accurately. This requirement means that Equation 28 must be met at all times.

Equation 28. AVSS + I_IDAC1 · (R_LEAD1 + R_RTD) + (I_IDAC1 + I_IDAC2) · (R_LEAD3 + R_REF) ≤ AVDD – 0.9 V

The device also offers the possibility to route the IDACs to the same inputs used for measurement. If the filter resistor values R_F1 and R_F2 are small enough and well matched, IDAC1 can be routed to AIN1 and IDAC2 to AIN0 in Figure 77. In this manner, even two 3-wire RTDs sharing the same reference resistor can be measured with a single device.

This design example discusses the implementation of a 3-wire Pt100 measurement to be used to measure temperatures ranging from –200°C to +850°C as stated in Table 23. The excitation current for the Pt100 is chosen as I_IDAC1 = 500 µA, which means a combined current of 1 mA is flowing through the reference resistor, R_REF. As mentioned previously, besides creating the reference voltage for the ADS1120, the voltage across R_REF also sets the common-mode voltage for the RTD measurement. In general, choose the largest reference voltage possible while still maintaining the compliance voltage of the IDACs as well as meeting the common-mode voltage requirement of the PGA. TI recommends setting the common-mode voltage at or near half the analog supply (in this case 3.3 V / 2 = 1.65 V), which in most cases satisfies the common-mode voltage requirements of the PGA. The value for R_REF is then calculated by Equation 29:

Equation 29. R_REF = V_ref / (I_IDAC1 + I_IDAC2) = 1.65 V / 1 mA = 1.65 kΩ

The stability of R_REF is critical to achieve good measurement accuracy over temperature and time. Choosing a reference resistor with a temperature coefficient of ±10 ppm/°C or better is advisable. If a 1.65 kΩ value is not readily available, another value near 1.65 kΩ (such as 1.62 kΩ or 1.69 kΩ) can certainly be used as well.

As a last step, the PGA gain must be selected in order to match the maximum input signal to the FSR of the ADC. The resistance of a Pt100 increases with temperature. Therefore, the maximum voltage to be measured (V_{IN (MAX)}) occurs at the positive temperature extreme. At 850°C, a Pt100 has an equivalent resistance of approximately 391 Ω as per the NIST tables. The voltage across the Pt100 equates to Equation 30:

Equation 30. V_{IN (MAX)} = V_{RTD (at 850°C)} = R_{RTD (at 850°C)} · I_IDAC1 = 391 Ω · 500 µA = 195.5 mV

The maximum gain that can be applied when using a 1.65-V reference is then calculated as (1.65 V / 195.5 mV) = 8.4. The next smaller PGA gain setting available in the ADS1120 is 8. At a gain of 8, the ADS1120 offers a FSR value as described in Equation 31:

Equation 31. FSR = ±V_ref / Gain = ±1.65 V / 8 = ±206.25 mV

This range allows for margin with respect to initial accuracy and drift of the IDACs and reference resistor.

After selecting the values for the IDACs, R_REF, and PGA gain, make sure to double check that the settings meet the common-mode voltage requirements of the PGA and the compliance voltage of the IDACs. To determine the true common-mode voltage at the ADC inputs (AIN0 and AIN1) the lead resistance must be taken into account as well.

The smallest common-mode voltage occurs at the lowest measurement temperature (–200°C) with R_LEADx = 0 Ω and is calculated using Equation 32 and Equation 33.

Equation 32. V_{CM (MIN)} = V_ref + (I_IDAC1 + I_IDAC2) · R_LEAD3 + I_IDAC2 · R_LEAD2 + ½ I_IDAC1 · R_{RTD (at –200°C)}

Equation 33. V_{CM (MIN)} = 1.65 V + ½ 500 µA · 18.52 Ω = 1.655 V

Actually, assuming V_{CM (MIN)} = V_ref is a sufficient approximation.

V_{CM (MIN)} must meet two requirements: Equation 15 requires V_{CM (MIN)} to be larger than AVDD / 4 = 3.3 V / 4 = 0.825 V and Equation 13 requires V_{CM (MIN)} to meet Equation 34:

Equation 34. V_{CM (MIN)} ≥ AVSS + 0.2 V + ½ Gain · V_{IN (MAX)} = 0 V + 0.2 V + ( ½ · 8 · 195.5 mV) = 982 mV

Both restrictions are satisfied in this design with a V_{CM (MIN)} = 1.65 V.

The largest common-mode voltage occurs at the highest measurement temperature (850°C) and is calculated using Equation 35 and Equation 36.

Equation 35. V_{CM (MAX)} = V_ref + (I_IDAC1 + I_IDAC2) · R_LEAD3 + I_IDAC2 · R_LEAD2 + ½ I_IDAC1 · R_{RTD (at 850°C)}

Equation 36. V_{CM (MAX)} = 1.65 V + 1 mA · 15 Ω + 500 µA · 15 Ω + ½ 500 µA · 391 Ω = 1.77 V

V_{CM (MAX)} does meet the requirement given by Equation 14, which in this design equates to Equation 37:

Equation 37. V_{CM (MAX)} ≤ AVDD – 0.2 V – ½ Gain · V_{IN (MAX)} = 3.3 V – 0.2 V – ( ½ · 8 · 195.5 mV) = 2.318 V

Finally, the maximum voltage that can occur on input AIN1 must be calculated to determine if the compliance voltage (AVDD – 0.9 V = 3.3 V – 0.9 V = 2.4 V) of IDAC1 is met. Note that the voltage on input AIN0 is smaller than the one on input AIN1. Equation 38 and Equation 39 show that the voltage on AIN1 is less than 2.4 V, even when taking the worst-case lead resistance into account.

Equation 38. V_{AIN1 (MAX)} = V_ref + (I_IDAC1 + I_IDAC2) · R_LEAD3 + I_IDAC1 · (R_{RTD (at 850°C)} + R_LEAD1)

Equation 39. V_{AIN1 (MAX)} = 1.65 V + 1 mA · 15 Ω + 500 µA · (391 Ω + 15 Ω) = 1.868 V

The register settings for this design are shown in Table 24.

Table 24. Register Settings

REGISTER	SETTING	DESCRIPTION
00h	66h	AINP = AIN1, AINN = AIN0, gain = 8, PGA enabled
01h	04h	DR = 20 SPS, normal mode, continuous conversion mode
02h	55h	External reference (REFP0, REFN0), simultaneous 50-Hz and 60-Hz rejection, IDAC = 500 µA
03h	70h	IDAC1 = AIN2, IDAC2 = AIN3

9.2.2.2.1 Design Variations for 2-Wire and 4-Wire RTD Measurements

Implementing a 2- or 4-wire RTD measurement is very similar to the 3-wire RTD measurement illustrated in Figure 77, except that only one IDAC is required.

Figure 78 shows a typical circuit implementation of a 2-wire RTD measurement. The main difference compared to a 3-wire RTD measurement is with respect to the lead resistance compensation. The voltage drop across the lead resistors, R_LEAD1 and R_LEAD2, in this configuration is directly part of the measurement (as shown in Equation 40) because there is no means to compensate the lead resistance by use of the second current source. Any compensation must be done by calibration.

Equation 40. V_IN = I_IDAC1 · (R_LEAD1 + R_RTD + R_LEAD2)

Figure 78. 2-Wire RTD Measurement

Figure 79 shows a typical circuit implementation of a 4-wire RTD measurement. Similar to the 2-wire RTD measurement, only one IDAC is required for exciting and measuring a 4-wire RTD in a ratiometric manner. The main benefit of using a 4-wire RTD is that the ADC inputs are connected to the RTD in the form of a Kelvin connection. Apart from the input leakage currents of the ADC, there is no current flow through the lead resistors R_LEAD2 and R_LEAD3 and therefore no voltage drop is created across them. The voltage at the ADC inputs consequently equals the voltage across the RTD and the lead resistance is of no concern.

Figure 79. 4-Wire RTD Measurement

Note that because only one IDAC is used and flows through the reference resistor, R_REF, the transfer function of a 2- and 4-wire RTD measurement differs compared to the one of a 3-wire RTD measurement by a factor of 2, as shown in Equation 41.

Equation 41. Code ∝ (R_{RTD (at Temperature)} · Gain) / R_REF

In addition, the common-mode and reference voltage is reduced compared to the 3-wire RTD configuration. Therefore, some further modifications may be required in case the 3-wire RTD design is used to measure 2- and 4-wire RTDs as well. If the decreased common-mode voltage does not meet the V_{CM (MIN)} requirements of the PGA anymore, either increase the value of R_REF by switching in a larger resistor or, alternatively, increase the excitation current while decreasing the gain at the same time.

9.2.2.3 Application Curves

Figure 80 and Figure 81 show the measurement results. The measurements are taken at T_A = 25°C. A system offset calibration is performed using a reference resistor of 100 Ω. No gain calibration is implemented. The data in Figure 80 are taken using precision resistors instead of a 3-wire Pt100. The respective temperature measurement error in Figure 81 is calculated from the data in Figure 80 using the NIST tables.

The design meets the required temperature measurement accuracy given in Table 23. Note that the measurement error shown in Figure 81 does not include the error of the RTD itself.

Figure 80. Resistance Measurement Error vs R_RTD

Figure 81. Temperature Measurement Error vs T_(RTD)

9.2.3 Resistive Bridge Measurement

The device offers several features to ease the implementation of ratiometric bridge measurements (such as a PGA with gains up to 128, buffered, differential reference inputs, and a low-side power switch).

Figure 82. Resistive Bridge Measurement