UCC25800-Q1 is an open-loop LLC transformer driver. By open-loop control, fixed output to input voltage gain can be achieved through the transformer turns ratio. The open-loop control also provides a simple design and superior noise immunity. The LLC topology utilizes the transformer leakage inductance as its resonant component, allowing the converter to use a transformer with a larger leakage inductance but a much smaller primary-side to secondary-side parasitic capacitance (inter-winding capacitance).

Other topologies, such as flyback or push-pull, require minimum leakage inductance to improve the efficiency, reduce the voltage and current stress, as well as minimize the noise created by the converter. In turn, this type of transformer suffers from larger inter-winding capacitance. When they are used in the gate driver bias supply applications, the high dv/dt from the inverter power stage can be coupled through the transformer inter-winding capacitance to the low-voltage side. This creates a much more severe EMI noise issue. It also puts more challenges on the common-mode transient immunity (CMTI). The open-loop LLC transformer driver solves this issue and provides a low-noise, robust solution for the isolated gate driver bias supplies.

Different from the typical PWM converters, LLC converters adjust the output voltage through varying its switching frequency. It is often called a PFM (Pulse Frequency Modulation) converter. As shown in Figure 2-1, the LLC converter has three resonant elements, the resonant inductor (L_r), the magnetizing inductor (L_m), and the resonant capacitor (C_r). In isolated bias supply designs, the transformer leakage inductance, and the magnetizing inductor can be used as part of the resonant circuit. The only external resonant component is the resonant capacitor.

The resonant frequency of the LLC converter is defined by the series resonance between the resonant inductor (L_r) and the resonant capacitor (C_r), as shown in Equation 1

When its switching frequency is equal or below the resonant frequency, the operation waveforms of LLC converter can be found in Figure 2-2.

When the switching frequency is equal to the resonant frequency, it can be observed that the transformer primary-side current is a sinusoidal shape. The secondary-side current is also a sinusoidal shape but with some phase shift to the primary side current. The phase shift is caused by the transformer magnetizing current. The output current is equal to the rectified transformer secondary-side current. In this operation mode, the resonant tank impedance is equal to zero, and the input and output voltages are connected virtually through the transformer. When the switching frequency moves away from the resonant frequency, the impedance of the resonant tank increases. As a result, the output voltage reduces. However, this only holds true when the switching frequency is higher than the resonant frequency, because the magnetizing inductor never participates the resonant and it doesn't influence the characteristic performance of the resonant circuit.

When the switching frequency is below the resonant frequency, it can be observed that the sinusoidal current becomes discontinuous. In the duration where the sinusoidal shape stops, transformer secondary-side current is equal to zero. During this period, the magnetizing inductor becomes part of the resonant circuit, but the resonant frequency is so low that the current shape appears linear. During this period, the magnetizing inductor stores more energy and transfers it to the secondary side through the resonant capacitor in the following half switching cycle. Therefore, the LLC is able to achieve higher voltage gain.

The LLC converter voltage gain refers to the relationship between its output voltage and its input voltage. It is defined as Equation 2. In this equation, n is the transformer primary side to secondary side turns ratio and the ½ comes from the half bridge configuration.

As shown in Figure 2-3, the voltage gain is affected by both switching frequency and load. In this set of curves, the switching frequency is normalized with the resonant frequency, which is defined in Equation 1, and Equation 3. The load can be normalized with the characteristic impedance and defined as Equation 4. Here, f_s is the switching frequency.

From this set of gain curves, at the resonant frequency, regardless of load conditions, the converter has a gain equal to 1, which means the relationship between the input and output voltage is only determined by the transformer turns ratio. This can be simply understood by the impedance of the resonant tank (L_r and C_r) is equal to zero at the resonant frequency. The input and output voltages are directly connected together virtually through the transformer.

If the LLC converter operates with a fixed switching frequency equal to the resonant frequency, the LLC converter is able to deliver a fixed voltage gain, with different load conditions. With a fixed input voltage, a fixed output voltage can be achieved.

In Figure 2-1, the resonant capacitor is on the primary side. From its gain curves, it can be observed when the switching frequency is below the resonant frequency, the converter voltage gain rises. This occurs because the energy that is stored in the magnetizing inductor is transferred to the secondary side in each half-switching cycle. The lower the switching frequency, the more energy is stored in the magnetizing inductor. Therefore the voltage gain keeps going up.

When the resonant capacitor is moved on the secondary side, the gain curves are changed. One example of using secondary-side resonant is shown in Figure 3-1.

Putting the resonant capacitor on the secondary side, when the switching frequency is below the resonant frequency, the energy stored in the magnetizing inductance can no longer be transferred to the secondary side. Instead, the energy is fed back to the input source. Due to this behavior, the voltage gain of the secondary-side resonant becomes flat when the switching frequency is below the resonant frequency, as shown in Figure 3-2.

Based on these curves, we can see if the LLC converter is operating with a fixed switching frequency and that frequency is slightly below the resonant frequency, the voltage gain is fixed, regardless of the frequency or the load condition, which means with a fixed input voltage, we get a fixed output voltage. This property also helps to maintain constant output voltage with the tolerances on the resonant components, which is equivalent to the switching frequency variation.

Besides the transformer turns-ratio, the LLC converter voltage gain is also affected by the rectification structure. In this section, a few rectification methods are presented and designers can select the one according to their cost structure from the components. Because the primary-side circuit remains the same, the description only focuses on the secondary-side circuit.

In this configuration, only one capacitor C_r is needed for the resonant capacitor. There is a DC voltage offset on the resonant capacitor and the voltage doubling is achieved through the transformer voltage in series with the DC offset of the resonant capacitor. When the transformer secondary-side voltage is positive, it adds together with the resonant capacitor offset voltage, diode D₁ is conducting, and the transformer delivers energy to the output. When the transformer voltage is negative, D₁ is off and D₂ is conducting. In this way, the output capacitor C_out is refreshed in one half of the switching cycle. In the other half switching cycle, the output capacitor gets no energy from the transformer and supplies the load using its stored energy. The output capacitor C_out should be much larger (> 10 times) than the resonant capacitor C_r. Or in other words, the output capacitor C_out can be considered as a high-frequency short at the switching frequency.

In this configuration, the resonant capacitor is split into two capacitors. The equivalent resonant capacitance is the sum of the two capacitances. Both of the resonant capacitors carry a DC offset equal to half of the output voltage. The output capacitor gets energy for each half of the switching cycle. The output capacitor C_out still needs to be much larger (> 10 times) than the resonant capacitor. Or in other words, the output capacitor C_out can be considered as a high-frequency short at the switching frequency

In this case, the full-wave rectifier is used. The output voltage gain is halved compared with the one using voltage doubler. The output capacitor C_out still needs to be much larger (> 10 times) than the resonant capacitor. Or in other words, the output capacitor C_out can be considered as a high-frequency short at the switching frequency

Other than these three rectification methods, the center-tap method can also be used. However, the center-tap method can only be used with primary side resonant and it is less preferred.

It can also be shown that, with all these three rectification methods, the transformer secondary side is in series with a capacitor. Together with the primary side DC blocking capacitor, the transformer saturation caused by DC offset can be avoided.

Table 4-1 summarizes the difference among these three rectification methods. In this table, "n" represents the transformer primary-side to secondary-side turns-ratio (N_P : N_S).

Given very few external components, the open-loop LLC converter design is mainly designing the LLC transformer. The LLC transformer design is a simple process that involves the selection of a suitable transformer turns ratio, volt-second rating, the transformer structure, and the AC resistance. The leakage inductance and magnetizing inductance would be the byproducts of the design, and the circuit can work with the parameters achieved.

When an LLC converter operates at its resonant frequency, the impedance of the resonant tank is equal to zero. The input voltage and output voltage are virtually connected together. Given this property, we can easily determine the transformer turns-ratio needed. Using the example shown in Figure 5-1, the primary side uses a half bridge. Therefore, the transformer primary side only sees half of the input voltage. The secondary side uses a voltage doubler (Here, the resonant capacitor is the sum of two capacitors ). Therefore, the transformer secondary side only sees half of the output voltage. In this case, considering the converter operates close to the resonant frequency, the voltage gain is equal to 1. We can get Equation 5.

Considering the voltage drop on the diodes, the transformer and the primary-side switches, we can set the transformer turns-ratio n using Equation 12. The extra 1 V can be adjusted as needed to account for the actual voltage drop or extra voltage needed for the post regulator.

Besides the transformer turns ratio, the volt-second (VS) rating is another important parameter. Given that the transformer primary side sees the half of the input voltage and the flux goes to both the first and third quadrant, the transformer volt-second rating should be calculated based on a quarter of the switching cycle and half of the input voltage. It can be determined by Equation 7. Here f_s is the switching frequency. Some design margin is recommended to consider the tolerance on the switching frequency, as well as the input voltage.

When the UCC25800-Q1 based bias supply is used in the inverter applications, especially for the high-side switches, the high dv/dt on the inverter switch node can couple through the bias supply transformer and causes extra EMI noise, as demonstrated in Table 5-1.

To minimize this noise coupling, it is desired to minimize the transformer primary-side to secondary-side capacitance. By physically distancing the primary-side winding and secondary-side winding, a transformer with split chamber bobbin can be used to achieve a minimum parasitic capacitance and simple manufacturing. The split chamber bobbin transformer is illustrated in Figure 5-2.

The transformer winding design starts with estimating the RMS currents of the primary-side and secondary-side windings . As discussed earlier, at the resonant frequency, the primary-side and secondary-side currents are both sinusoidal. There is a phase shift between these two currents, due to the magnetizing current. If the magnetizing current is ignored, the primary-side current and secondary-side current are scaled by the transformer turns-ratio. Since the output current is equal to the average value of the rectified transformer secondary-side current, the RMS current of the transformer windings can be easily estimated by Equation 8 and Equation 9. Considering the tolerances on the switching frequency and component values, an extra 20~30% design margin is recommended for these currents.

Secondary-side winding RMS current:

Primary-side winding RMS current:

Even though the converter gain is equal to the transformer turns-ratio when the switching frequency is equal to the resonant frequency, the output voltage will be slightly lower than the theoretical value due to the loss elements in the converter, including the diode voltage drop (V_f), the primary-side switch on-state resistance (R_dson), the transformer AC resistance R_ac, the resonant capacitor ESR (R_ESR), as well as the diode ESR (R_Diode). The overall output voltage can be estimated based on Equation 10. Given the transformer AC resistance affects the voltage gain, to improve the load regulation, it is desired to minimize the transformer AC resistance.

The transformer AC resistance should be measured from the secondary side, with primary side shorted, and at the resonant frequency, as show in Figure 5-3.

Ideally, this impedance should be as low as possible. Based on the equation, we can estimate the voltage drop caused by the load current and estimate how much AC resistance can be allocated for the transformer. The R_dson is coming from UCC25800-Q1 and it can be estimated using 0.3 Ω. The ESR of the capacitor can be ignored if the NP0 capacitor is used. The X7R capacitor would have a larger ESR. The diode resistance can be estimated based on the diode forward voltage drop curves, normally it is about 0.3 Ω.

As described earlier, the transformer parasitic inductance, including the leakage inductance and the magnetizing inductance can be used as part of the resonant circuit. The resonant capacitor can be designed based on these inductances to match the desired switching frequency.

When the split chamber bobbin is used, the primary-side winding and secondary-side winding are physically separated. The coupling between two windings is poor and large leakage inductance is created. When using the secondary-side resonant, the method described in Figure 5-3 can be used to measure the leakage inductance (L_k) from the transformer secondary side. The leakage inductance is measured on the secondary side when the primary side is shorted. Following the same principle, when the primary-side resonant is used, measure the leakage inductance from the transformer primary side with the secondary side shorted. Once the leakage inductance is measured, the resonant capacitor C_r can be selected according to Equation 11. In this equation, L_k is the leakage inductance measured and f₀ is the resonant frequency. The resonant frequency can be chosen as 10% above the switching frequency to allow component tolerances. If the one resonant capacitor configuration is used, the resonant capacitor should be the calculated total resonant capacitor C_r.