• 符合汽车应用要求
• 具有符合 AEC-Q100 标准的下列特性：
  • 器件温度等级 1：–40°C 至 +125°C 的工作环境温度范围
  • 器件 HBM ESD 分类等级 2
  • 器件 CDM ESD 分类等级 C6
• 提供功能安全
  • 有助于进行功能安全系统设计的文档
• 宽共模输入范围：0V 至 36V
• 双输出：放大器和比较器输出
• 高精度放大器：
  • 失调电压：35 µV（最大值）
  • 失调电压漂移：0.5 µV/°C（最大值）
  • 增益误差：0.1%（最大值）
  • 增益误差漂移：10 ppm/°C
• 可用放大器增益：
  • INA301A1-Q1：20 V/V
  • INA301A2-Q1：50 V/V
  • INA301A3-Q1：100 V/V
• 可编程警报阈值，通过单个电阻器设置
• 总警报响应时间：1 µs
• 透明模式和锁存模式下的开漏输出
• 封装：VSSOP-8

• 电磁阀控制
• 低侧电机监控
• 电子动力转向
• 电动座椅
• 电动车窗
• 车身控制模块
• 电子控制单元
• 过流保护
• 电子保险丝

INA301-Q1 由高共模电流感测放大器和高速比较器组成，通过测量电流感测或分流电阻两侧的电压并将该电压与定义的阈值限值相比较来提供过流保护。此器件具有一个可调限制阈值范围，此范围由单个外部限值设定电阻器设置。该分流监控器能够在 0V 至 36V 的共模电压范围内测量差分电压信号，与电源电压无关。

开漏报警输出可配置为透明模式（输出状态与输入状态保持一致）或锁存模式（复位锁存时清除报警输出）。器件报警响应时间不到 1 µs，能够快速检测过流事件。

这款器件由 2.7V-5.5V 单电源供电运行，汲取的最大电源电流为 700 µA。该器件在 -40°C 至 +125°C 的扩展级温度范围下额定运行，并且采用 8 引脚 VSSOP 封装。

典型应用

at T_A = 25°C, V_S = 5 V, V_IN+ = 12 V, and alert pullup resistor = 10 kΩ (unless otherwise noted)

Figure 6-1 Input Offset Voltage Distribution (INA301A1-Q1)

Figure 6-3 Input Offset Voltage Distribution (INA301A3-Q1)

Figure 6-5 Common-Mode Rejection Ratio Distribution (INA301A1-Q1)

Figure 6-7 Common-Mode Rejection Ratio Distribution (INA301A3-Q1)

Figure 6-9 Common-Mode Rejection Ratio vs. Frequency

Figure 6-11 Gain Error Distribution (INA301A2-Q1)

Figure 6-13 Gain Error vs. Temperature

Figure 6-15 Power-Supply Rejection Ratio vs. Frequency

VS = 5 V

Figure 6-17 Input Bias Current vs. Common-Mode Voltage

Figure 6-19 Input Bias Current vs. Temperature

Figure 6-21 Quiescent Current vs. Temperature

Figure 6-23 0.1-Hz to 10-Hz Referred-to-Input Voltage Noise

4-VPP output step

Figure 6-25 Voltage Output Falling Step Response

Figure 6-27 Start-Up Response

Figure 6-29 Total Propagation Delay (INA301A1-Q1)

Figure 6-31 Total Propagation Delay (INA301A3-Q1)

Figure 6-33 Comparator Alert V_OL vs. I_OL

Figure 6-35 Comparator Reset Response

Figure 6-2 Input Offset Voltage Distribution (INA301A2-Q1)

Figure 6-4 Input Offset Voltage vs. Temperature

Figure 6-6 Common-Mode Rejection Ratio Distribution (INA301A2-Q1)

Figure 6-8 Common-Mode Rejection Ratio vs. Temperature

Figure 6-10 Gain Error Distribution (INA301A1-Q1)

Figure 6-12 Gain Error Distribution (INA301A3-Q1)

Figure 6-14 Gain vs. Frequency

Figure 6-16 Output Voltage Swing vs. Output Current

VS = 0 V

Figure 6-18 Input Bias Current vs. Common-Mode Voltage

Figure 6-20 Quiescent Current vs. Supply Voltage

Figure 6-22 Input-Referred Voltage Noise vs. Frequency

4-VPP output step

Figure 6-24 Voltage Output Rising Step Response

Figure 6-26 Common-Mode Voltage Transient Response

Figure 6-28 Limit Current Source vs. Temperature

Figure 6-30 Total Propagation Delay (INA301A2-Q1)

VOD = 1 mV

Figure 6-32 Comparator Propagation Delay vs. Temperature

Figure 6-34 Hysteresis vs. Temperature

The INA301-Q1 is a 36-V common-mode, zero-drift topology, current-sensing amplifier that can be used in both low-side and high-side configurations. These specially-designed, current-sensing amplifiers are able to accurately measure voltages developed across current-sensing resistors (also known as current-shunt resistors) on common-mode voltages that far exceed the supply voltage powering the device. Current can be measured on input voltage rails as high as 36 V, and the device can be powered from supply voltages as low as 2.7 V. The device can also withstand the full 36-V common-mode voltage at the input pins when the supply voltage is removed without causing damage.

The zero-drift topology enables high-precision measurements with maximum input offset voltages as low as 35 μV with a temperature contribution of only 0.5 μV/°C over the full temperature range of –40°C to +125°C. The low total offset voltage of the INA301-Q1 enables smaller current-sense resistor values to be used, and allows for a more efficient system operation without sacrificing measurement accuracy resulting from the smaller input signal.

The INA301-Q1 uses a single external resistor to allow for a simple method of setting the corresponding current threshold level for the device to use for out-of-range comparison. Combining the precision measurement of the current-sense amplifier and the onboard comparator enables an all-in-one overcurrent detection device. This combination creates a highly-accurate solution that is capable of fast detection of out-of-range conditions, and allows the system to take corrective actions to prevent potential component or system-wide damage.

The device ALERT pin is an active-low, open-drain output that is designed to be pulled low when the input conditions are detected to be out-of-range. Add a 10-kΩ pullup resistor from ALERT pin to the supply voltage. This open-drain pin can be pulled up to a voltage beyond the V_S supply voltage, but must not exceed 5.5 V.

Figure 7-1 shows the alert output response of the internal comparator. When the output voltage of the amplifier is less than the voltage developed at the LIMIT pin, the comparator output is in the default high state. When the amplifier output voltage exceeds the threshold voltage set at the LIMIT pin, the comparator output becomes active and pulls low. This active low output indicates that the measured signal at the amplifier input has exceeded the programmed threshold level, indicating an overcurrent or out-of-range condition has occurred.

Figure 7-1 Overcurrent Alert Response

The INA301-Q1 determines if an overcurrent event is present by comparing the amplified measured voltage developed across the current-sensing resistor to the corresponding signal developed at the LIMIT pin. The threshold voltage for the LIMIT pin is set using a single external resistor, or by connecting an external voltage source to the LIMIT pin.

The typical method for setting the limit threshold voltage is to connect a resistor from the LIMIT pin to ground. The value of this resistor, R_LIMIT, is chosen in order to create a corresponding voltage at the LIMIT pin equivalent to the output voltage, V_OUT, when the maximum desired load current is flowing through the current-sensing resistor. An internal 80-µA current source is connected to the LIMIT pin to create a corresponding voltage used to compare to the amplifier output voltage, depending on the value of the R_LIMIT resistor.

In the equations from Table 7-1, V_TRIP represents the overcurrent threshold that the device is programmed to monitor, and V_LIMIT is the programmed signal set to detect the V_TRIP level.

If the current level indicating an out-of-range condition is present is 20 A, and the current-sense resistor value is 10 mΩ, then the input threshold signal is 200 mV. The INA301A1-Q1 has a gain of 20, therefore, the resulting output voltage at the 20-A input condition is 4 V. The value for R_LIMIT is selected to allow the device to detect to this 20-A threshold, indicating an overcurrent event occurred. When the INA301-Q1 detects this out-of-range condition, the ALERT pin asserts and pulls low. For this example, Table 7-2 lists the calculated value of R_LIMIT required to detect a 4-V level as 50 kΩ.

Another method for setting the limit voltage is to connect the LIMIT pin to a programmable digital-to-analog converter (DAC) or other external voltage source. The benefit of this method is the ability to adjust the current-limit threshold to account for different threshold voltages that are used for different system operating conditions. For example, this method can be used in a system that has one current-limit threshold level that must be monitored during a power-up sequence, but different threshold levels that must be monitored during other system operating modes.

In Table 7-3, V_TRIP represents the overcurrent threshold that the device is programmed to monitor, and V_SOURCE is the programmed signal set to detect the V_TRIP level.

The onboard comparator in the INA301-Q1 reduces the possibility of oscillations in the alert output when the measured signal level is near the overlimit threshold level because of noise. When the output voltage (V_OUT) exceeds the voltage developed at the LIMIT pin, the ALERT pin is asserted and pulls low. The output voltage must drop below the LIMIT pin threshold voltage by the gain-dependent hysteresis level for the ALERT pin to deassert and return to the nominal high state (see Figure 7-2).

Figure 7-2 Typical Comparator Hysteresis

The device has two output operating modes, transparent and latched, that are selected based on the RESET pin setting. These modes change how the ALERT pin responds following an alert when the overcurrent condition is removed.

The device is set to transparent mode when the RESET pin is pulled low, thus allowing the output alert state to change and follow the input signal with respect to the programmed alert threshold. For example, when the differential input signal rises above the alert threshold, the ALERT output pin is pulled low. As soon as the differential input signal drops below the alert threshold, the output returns to the default high-output state. A common implementation using the device in transparent mode is to connect the ALERT pin to a hardware interrupt input on a microcontroller. As soon as an overcurrent condition is detected and the ALERT pin is pulled low, the hardware interrupt input detects the output-state change, and the microcontroller can begin to make changes to the system operation required to address the overcurrent condition. Under this configuration, the ALERT pin transition from high to low is captured by the microcontroller so that the output can return to the default high state when the overcurrent event is removed.

Some applications do not have the functionality available to continuously monitor the state of the output ALERT pin to detect an overcurrent condition as described in the Section 7.4.1.1 section. A typical example of this application is a system that is only able to poll the ALERT pin state periodically to determine if the system is functioning correctly. If the device is set to transparent mode in this type of application, the state change of the ALERT pin might be missed when ALERT is pulled low to indicate an out-of-range event, if the out-of-range condition does not appear during one of these periodic polling events. Latch mode is specifically intended to accommodate these applications.

The INA301-Q1 is placed into the corresponding output modes based on the signal connected to RESET (see Table 7-4). The difference between latch mode and transparent mode is how the ALERT pin responds when an overcurrent event ends. In transparent mode (RESET = low), when the differential input signal drops below the limit threshold level after the ALERT pin asserts because of an overcurrent event, the ALERT pin state returns to the default high setting to indicate that the overcurrent event has ended.

In latch mode (RESET = high), when an overlimit condition is detected and the ALERT pin is pulled low, the ALERT pin does not return to the default high state when the differential input signal drops below the alert threshold level. In order to clear the alert, pull the RESET pin low for at least 100 ns. Pulling the RESET pin low allows the ALERT pin to return to the default high level, provided that the differential input signal has dropped below the alert threshold. If the input signal is still greater than the threshold limit when the RESET pin is pulled low, the ALERT pin remains low. When the alert condition is detected by the system controller, the RESET pin can be set back to high in order to place the device back in latch mode.

The latch and transparent modes represented in Figure 7-3 show that when V_IN drops back below the V_LIMIT threshold for the first time, the RESET pin is pulled high. With the RESET pin is pulled high, the device is set to latch mode, so that the ALERT pin output state does not return high when the input signal drops below the V_LIMIT threshold. Only when the RESET pin is pulled low does the ALERT pin return to the default high level, thus indicating that the input signal is below the limit threshold. When the input signal drops below the limit threshold for the second time, the RESET pin is already pulled low. The device is set to transparent mode at this point and the ALERT pin is pulled back high as soon as the input signal drops below the alert threshold.

Figure 7-3 Transparent Mode vs. Latch Mode

The INA301-Q1 enables easy configuration to detect overcurrent conditions in an application. This device is individually targeted towards unidirectional overcurrent detection of a single threshold. However, this device can also be paired with additional INA301-Q1 devices and circuitry to create more complex monitoring functional blocks.

The INA301-Q1 measures the differential voltage developed across a resistor when current flows through the component in order to determine if the current being monitored exceeds a defined limit. This resistor is commonly referred to as a current-sensing resistor or a current-shunt resistor, with each term commonly used interchangeably. The flexible design of this device allows for measuring a wide differential input signal range across the current-sensing resistor.

Selecting the value of this current-sensing resistor is primarily based on two factors: the required accuracy of the current measurement, and the allowable power dissipation across the current-sensing resistor. Larger voltages developed across this resistor allow for more accurate measurements to be made. Amplifiers have fixed internal errors that are largely dominated by the inherent input offset voltage. When the input signal decreases, these fixed internal amplifier errors become a larger portion of the measurement and increase the uncertainty in the measurement accuracy. When the input signal increases, the measurement uncertainty is reduced because the fixed errors are a smaller percentage of the signal being measured. Therefore, the use of larger-value, current-sensing resistors inherently improves measurement accuracy.

However, a system design trade-off must be evaluated through the use of larger input signals that improve measurement accuracy. Increasing the current sense resistor value results in an increase in power dissipation across the current-sensing resistor, and also increases the differential voltage developed across the resistor when current passes through the component. This increase in voltage across the resistor increases the power that the resistor must be able to dissipate. Decreasing the value of the current-shunt resistor reduces the power dissipation requirements of the resistor, but increases the measurement errors resulting from the decreased input signal. Selecting the optimal value for the shunt resistor requires factoring both the accuracy requirement for the specific application, and the allowable power dissipation of this component.

Low-ohmic-value resistors enable large currents to be accurately monitored with the INA301-Q1. An increasing number of very low-ohmic-value resistors are becoming more widely available, with values of 200 μΩ and less, and power dissipations of up to 5 W.

In this example, the trade-offs involved in selecting a current-sensing resistor are described. This example requires 2.5% accuracy for detecting a 10-A overcurrent event, with only 250 mW of allowable power dissipation across the current-sensing resistor at the full-scale current level. Although the maximum power dissipation is defined as 250 mW, a lower dissipation is preferred in order to improve system efficiency. Some initial assumptions are made that are used in this example:

• the limit-setting resistor (R_LIMIT) is a 1% component
• the maximum tolerance specification for the internal threshold setting current source (0.5%) is used

Given the total error budget of 2.5%, up to 1% of error is available to be attributed to the measurement error of the device under these conditions.

As shown in Table 8-1, the maximum value calculated for the current-sensing resistor with these requirements is 2.5 mΩ. Although this value satisfies the maximum power dissipation requirement of 250 mW, headroom is available from the 2.5% maximum total overcurrent detection error in order to reduce the value of the current-sensing resistor, and reduce the power dissipation further. Selecting a 1.5-mΩ, current-sensing resistor value offers a good tradeoff for reducing the power dissipation in this scenario by approximately 40% while still remaining within the accuracy region.

External system noise can significantly affect the ability of a comparator to accurately measure and detect whether input signals exceed the reference threshold levels and reliably indicate overrange conditions. The most obvious effect that external noise has on the operation of a comparator is to cause a false-alert condition. If a comparator detects a large noise transient coupled into the signal, the device can easily interpret this transient as an overrange condition.

External filtering helps reduce the amount of noise that reaches the comparator, and thus reduce the likelihood of a false alert from occurring. The tradeoff to adding this noise filter is that the alert response time is increased because of the input signal being filtered along with the noise. Figure 8-1 shows the implementation of an input filter for the device.

Figure 8-1 Input Filter

Limiting the input resistance this filter is important because this resistance can have a significant affect on the input signal that reaches the device input pins because of the device input bias currents. A typical system implementation involves placing the current-sensing resistor very near the device so that the traces are very short and the trace impedance is very small. This layout helps reduce the ability of coupling additional noise into the measurement. Under these conditions, the characteristics of the input bias currents have minimal affect on device performance.

As illustrated in Figure 8-2, the input bias currents increase in opposite directions when the differential input voltage increases. This increase results from a device design that allows common-mode input voltages to far exceed the device supply voltage range. With input filter resistors now placed in series with these unequal input bias currents, there are unequal voltage drops developed across these input resistors. The difference between these two voltage drops appears as an added signal that, in this case, subtracts from the voltage developed across the current-sensing resistor, thus reducing the signal that reaches the device input pins. Smaller-value input resistors reduce this effect of signal attenuation to allow for a more accurate measurement.

Figure 8-2 Input Bias Current vs. Differential Input Voltage

For example, with a differential voltage of 10 mV developed across a current-sensing resistor and using 20-Ω resistors, the differential signal that actually reaches the device is 9.85 mV. A measurement error of 1.5% is created as a result of these external input filter resistors. Use 10-Ω input filter resistors instead of the 20-Ω resistors to reduce this added error from 1.5% down to 0.75%.

With a small amount of additional circuitry, the INA301-Q1 can be used in circuits subject to transients greater than 36 V. Use only Zener diodes or Zener-type transient absorbers (sometimes referred to as transzorbs). Any other type of transient absorber has an unacceptable time delay. Start by adding a pair of resistors as a working impedance for the Zener diode, as shown in Figure 8-3. Keep these resistors as small as possible; preferably, 10 Ω or less. Larger values can be used, but with an additional induced error resulting from less signal reaching the device input pins. Because this circuit limits only short-term transients, many applications are satisfied with a 10-Ω resistor along with conventional Zener diodes of the lowest power rating available. This combination uses the least amount of board space. These diodes can be found in packages as small as SOT-523 or SOD-523.

Figure 8-3 Transient Protection

Although this device is only able to measure current through a current-sensing resistor flowing in one direction, a second INA301-Q1 can be used to create a bidirectional monitor (see Figure 8-4).

Figure 8-4 Bidirectional Application

For this design example, use the parameters listed in Table 8-2 as the input parameters.

First, reverse the input pins of the second INA301-Q1 across the current-sensing resistor. The second device is now able to detect current flowing in the other direction relative to the first device.

Then, select limit resistors to set the voltage trip points by using the equations in Table 7-1. For this application example, these equations give a value of 6.25 kΩ for both limit resistors.

Connect the outputs of each device to an AND gate in order to detect if either of the limit threshold levels are exceeded. Table 8-3shows that the output of the AND gate is high if neither overcurrent limit thresholds are exceeded. A low output state of the AND gate indicates that either the positive overcurrent limit or the negative overcurrent limit are surpassed.

Figure 8-5 shows two INA301-Q1 devices being used in a bidirectional configuration and an output control circuit to detect if one of the two alerts is exceeded.

Figure 8-5 Bidirectional Application Curve

The device input circuitry accurately measures signals on common-mode voltages beyond the power-supply voltage, V_S. For example, the voltage applied to the VS power-supply pin can be 5 V, whereas the load power-supply voltage being monitored (V_CM) can be as high as 36 V. At power up, for applications where the common-mode voltage (V_CM) slew rate is greater than 6 V/μs with a final common-mode voltage greater than 20 V, TI recommends that the V_S supply be present before V_CM. If the use case requires V_CM to be present before V_S with V_CM under these same slewing conditions, then a 331-Ω resistor must be added between the V_S supply and the V_S pin bypass capacitor.

Power-supply bypass capacitors are required for stability and must be placed as close as possible to the supply and ground pins of the device. A typical value for this supply bypass capacitor is 0.1 µF. Applications with noisy or high-impedance power supplies may require additional decoupling capacitors to reject power-supply noise.

During slow power-up events, current flow through the sense resistor or voltage applied to the REF pin can result in the output voltage momentarily exceeding the voltage at the LIMITx pins, resulting in an erroneous indication of an out-of-range event on the ALERTx output. When powering the device with a slow ramping power rail where an input signal is already present, all alert indications should be disregarded until the supply voltage has reached the final value.