8.3.1 Wideband Current Feedback Operation

The OPA695 provides a new level of performance in wideband current feedback operational amplifiers. Nearly constant AC performance over a wide gain range, along with 4300-V/μs slew rate, gives a lower power and cost solution for high-intercept IF amplifier requirements. While optimized at a gain of +8 V/V (12 dB to a matched 50-Ω load) to give 450-MHz bandwidth, applications from gains of 1 to 40 can be supported. As a gain of +2 video line driver, the bandwidth extends to 1.4 GHz with a slew rate to support the highest pixel rates. At gains above 20, the signal bandwidth starts to decrease, but still exceeds 180 MHz up to a gain of 40 V/V (26 dB to a matched 50-Ω load). Single +5-V supply operation is also supported with similar bandwidths but reduced output power capability. For lower speed (< 250-MHz) requirements with higher output powers, consider the OPA691.

Figure 48 shows the DC-coupled, gain of +8 V/V, dual-power supply circuit used as the basis of the ±5-V Specifications and Typical Characteristic curves. For test purposes, the input impedance is set to 50 Ω with a resistor to ground, and the output impedance is set to 50 Ω with a series output resistor. Voltage swings reported in the specifications are taken directly at the input and output pins, while load powers (dBm) are defined at a matched 50-Ω load. For the circuit of Figure 48, the total effective load is 100 Ω || 458 Ω = 82 Ω. The disable control line (DIS) is typically left open for normal amplifier operation. The disable line must be asserted low to shut off the OPA695. One optional component is included in Figure 48. In addition to the usual power supply decoupling capacitors to ground, a 0.01-μF capacitor is included between the two power supply pins. In practical PCB layouts, this optional added capacitor typically improves the 2nd-harmonic distortion performance by 3 dB to 6 dB for bipolar supply operation.

Figure 49 shows the DC-coupled, gain of –8 V/V, dual-power supply circuit used as the basis of the Inverting Typical Characteristic curves. Inverting operation offers several performance benefits. Because there is no common-mode signal across the input stage, the slew rate for inverting operation is higher and the distortion performance is slightly improved. An additional input resistor, R_T, is included in Figure 49 to set the input impedance equal to 50 Ω. The parallel combination of R_T and R_G set the input impedance. Both the non-inverting and inverting applications of Figure 48 and Figure 49 benefit from optimizing the feedback resistor (R_F) value for bandwidth (see the discussion in Setting Resistor Values to Optimize Bandwidth). The typical design sequence is to select the R_F value for best bandwidth, set R_G for the gain, then set R_T for the desired input impedance. As the gain increases for the inverting configuration, a point is reached where R_G equals 50 Ω, where R_T is removed, and the input match is set by R_G only. With R_G fixed to achieve an input match to 50 Ω, R_F is increased to increase gain. This quickly reduces the achievable bandwidth, as shown by the inverting gain of –16 frequency response in the Typical Characteristic curves. For gains > 10 V/V (14 dB at the matched load), noninverting operation is recommended to maintain broader bandwidth.

Figure 48. DC-Coupled, G = +8 V/V, Bipolar Supply Specifications and Test Circuit

Figure 49. DC-Coupled, G = –8 V/V, Bipolar Supply Specifications and Test Circuit

Figure 50 shows the AC-coupled, single +5-V supply, gain of +8 V/V circuit configuration used as a basis for the +5V-only Specifications and Typical Characteristic curves. The key requirement for broadband single-supply operation is to maintain input and output signal swings within the useable voltage ranges at both the input and the output. The circuit of Figure 50 establishes an input midpoint bias using a simple resistive divider from the +5-V supply (two 806-Ω resistors) to the noninverting input. The input signal is then AC-coupled into this midpoint-voltage bias. The input voltage can swing to within 1.6 V of either supply pin, giving a 1.8-V_PP input signal range centered between the supply pins. The input impedance matching resistor (57.6 Ω) used in Figure 50 is adjusted to give a 50-Ω input match when the parallel combination of the biasing divider network is included. The gain resistor (R_G) is AC-coupled, giving the circuit a DC gain of +1. This puts the input DC bias voltage (2.5 V) on the output as well. The feedback resistor value has been adjusted from the bipolar supply condition to re-optimize for a flat frequency response in +5 V only, gain of +8 operation (see Setting Resistor Values to Optimize Bandwidth). On a single +5-V supply, the output voltage can swing to within 1.0 V of either supply pin while delivering more than 90-mA output current, giving 3-V output swing into 100 Ω (7-dBm maximum at the matched load). The circuit in Figure 50 shows a blocking capacitor driving into a 50-Ω output resistor, then into a 50-Ω load. Alternatively, the blocking capacitor could be removed with the load tied to a supply midpoint, or to ground if the DC current required by this grounded load is acceptable.

Figure 51 shows the AC-coupled, single +5-V supply, gain of –8 V/V circuit configuration used as a basis for the +5V-only Typical Characteristic curves. In this case, the midpoint DC bias on the noninverting input is also de-coupled with an additional 0.1-μF decoupling capacitor. This reduces the source impedance at higher frequencies for the noninverting input bias current noise. This 2.5-V bias on the noninverting input pin appears on the inverting input pin and, because R_G is DC-blocked by the input capacitor, also appears at the output pin. One advantage to inverting operation is that as there is no signal swing across the input stage, higher slew rates and operation to lower supply voltages are possible. To retain a 1-V_PP output capability, operation down to a 3-V supply is allowed. At a +3-V supply, the input common mode range is 0 V. However, for the inverting configuration of a current feedback amplifier, wideband operation is retained even with the input stage saturated.

Figure 50. AC-Coupled, G = +8 V/V, Single-Supply Specifications and Test Circuit

Figure 51. AC-Coupled, G = –8 V/V, Single-Supply Specifications and Test Circuit

The single-supply test circuits of Figure 50 and Figure 51 show +5-V operation. These same circuits can be used over a single-supply range of +5 V to +12 V. Operating on a single +12-V supply, with the Absolute Maximum Supply voltage specification of +13 V, gives adequate design margin for the typical ±5% supply tolerance.

8.3.2 RF Specifications and Applications

The ultra-high, full-power bandwidth and 3rd-order intercept of the OPA695 are ideal for IF amplifier applications. The advantage of a wideband operational amplifier such as the OPA695 include good (and independent) I/O impedance matching, as well as high reverse isolation. A designer accustomed to fixed-gain RF amplifiers will get almost perfect gain accuracy, higher I/O return loss, and 3rd-order intercept points exceeding 30 dBm (up to 110 MHz) using only a 13-mA supply current for the OPA695. Using the considerable design freedom achieved by adjusting the external resistors, the OPA695 can replace a wide range of fixed-gain RF amplifiers with a single part. To understand (in RF amplifier terms) how to take advantage of this, consider first the 4-S parameters (see the example circuits of Figure 48 and Figure 49 on ±5-V supplies, but similar results can be obtained on a single +5-V to +12-V supply).

8.3.3 Input Return Loss (S₁₁)

Input return loss is a measure of how closely (over frequency) the input impedance matches the source impedance. This is relatively independent of gain setting for both the noninverting and inverting configurations. The Typical Characteristics show the magnitude of S₁₁ for the circuits of Figure 48 and Figure 49 through 1 GHz (noninverting gain of +8 and inverting gain of –8 operation, respectively). Noninverting operation offers much better matching to higher frequencies, with the only deviation due to the parasitic input capacitance of the input pin. The noninverting input match is simply set by the resistor to ground on the noninverting input, as the amplifier itself shows a very high input impedance. Inverting operation is also good, but rises more quickly due to loop gain roll-off effects appearing at the inverting node. The inverting mode input match is set by the parallel combination of R_G and R_T in Figure 49, as the inverting amplifier node may be considered a virtual ground. A good, fixed-gain, RF amplifier would have an input, Voltage Standing Wave Ratio (VSWR) < 1.2:1. This corresponds to an S₁₁ of –21 dB. The OPA695 exceeds this performance through 100 MHz for the inverting mode of operation, and through 400 MHz for the noninverting mode.

8.3.4 Output Return Loss (S₂₂)

Output return loss is a measure of how closely (over frequency) the output impedance matches the load impedance. This is relatively independent of gain setting for both the noninverting and inverting configurations. The output matching impedance, to a first order, is set by adding a series resistor to the low impedance output of the operational amplifier.

Because the operational amplifier itself shows a low output impedance that increases with frequency, an improvement in the output match can therefore be obtained by adding a small equalizing capacitor across this output resistor. The Typical Characteristics show the measured S₂₂ with and without this 2.5-pF capacitor (across the 50-Ω output resistor). Again, a good match for a fixed-gain RF amplifier would give a VSWR of 1.2:1 (S₂₂ < –21 dB). The Typical Characteristic curves show that a simple 50-Ω output resistor holds better than –21 dB to 140 MHz, but up to 380 MHz with the tuning capacitor.

8.3.5 Forward Gain (S₂₁)

In all high-speed amplifier data sheets, forward gain is the small signal gain plotted over frequency. The difference between noninverting and inverting operation is that the phase of S₂₁ starts out at 0° for the noninverting and –180° for the inverting. This initial phase shift for inverting mode is inconsequential to most IF strip applications. The phase of S₂₁ was not shown in Typical Characteristics, but is linear with frequency and may be accurately modeled as a constant time delay through the amplifier.

The Typical Characteristics show S₂₁ over a range of signal gains, where the external resistors have been adjusted to re-optimize flatness at each gain setting. Because this is a current feedback operational amplifier, the signal bandwidth can be held relatively constant as the desired gain setting is changed. The plot of the noninverting bandwidth versus gain shows some change in bandwidth versus gain (due to parasitic capacitive effects on the inverting node) with very little change showing up for the inverting mode of operation.

Signal gains are most often referred to as V/V in operational amplifier data sheets. This is the voltage gain from input to output and is set by external resistor ratios. Because the output impedance is set by a physical series resistor, the voltage gain to the matched load is cut in half by this resistor divider. The log gain to the matched load for the noninverting circuit of Figure 48 is:

Equation 1.

The log gain to the matched load for the inverting circuit of Figure 49 is:

Equation 2.

The specific resistor values used in Figure 48 and Figure 49 give both a maximally-flat bandwidth and a 12-dB gain to the matched load. The design tables located in the Noise Figure section summarize the required resistor values over a range of desired gains for the circuits of Figure 48 and Figure 49.

As the desired signal gain increases, the achievable bandwidths decrease. In the noninverting case, it decreases relatively quickly as shown in Typical Characteristics. The inverting configuration holds almost constant bandwidth (with correctly selected external resistor values) until R_G reduces to equal 50 Ω, and remains at that value to satisfy the input impedance matching requirement, with further increases in gain achieved by increasing R_F in Figure 49. The bandwidth then decreases rapidly as shown by the gain of –16 V/V plot in Typical Characteristics.

8.3.6 Reverse Isolation (S₁₂)

Reverse isolation is a measure of how much power injected into the output pin returns to the source. This is rarely specified for an operational amplifier because operational amplifiers are nearly uni-directional signal devices. Below 300 MHz, the noninverting configuration of Figure 48 gives much better isolation than the inverting of Figure 49. Both are well below 40-dB isolation through 350 MHz.

8.3.7 Limits to Dynamic Range

The next set of considerations for RF amplifier applications are the defined limits to dynamic range. Typical fixed-gain RF amplifiers include:

• –1-dB compression (a measure of maximum output power)
• Two-tone, 3rd-order, output intermodulation intercept (a measure of achievable spurious-free dynamic range)
• Noise figure (a measure of degradation in signal to noise ratio in passing through the amplifier)

8.3.7.1 –1-dB Compression

The definition for –1-dB compression power is output power where the actual power is 1 dB less than the input power, plus the log gain. In classic RF amplifiers, this is typically 10 dB less than the 3rd-order intercept. That relationship does not hold for operational amplifiers, as their intercept is improved by loop gain to be far more than 10 dB higher than the –1-dB compression. A simple estimate for –1-dB compression for the OPA695 is the maximum non-slew limited output voltage swing available at the matched load, converted into a power with 1 dB added to satisfy the definition. For the OPA695 on ±5-V supplies, its output will deliver approximately ±4.0 V at the output pin or ±2.0 V at the matched load. The conversion from V_PP to power (for a sine wave) is:

Equation 3.

Converting this 4.0-V_PP swing at the load to dBm gives 16 dBm; adding 1 dB to this (to satisfy the definition) gives a –1-dB compression of 17 dBm for the OPA695 operating on ±5-V supplies. This is a good estimate for frequencies that require less than the full slew rate of the OPA695.

The maximum frequency of operation given an available slew rate and desired peak output swing (at the output pin for a sine wave) is:

Equation 4.

Putting in the 4600-V/μs slew rate available in the inverting mode of operation and the 4.0-V peak output swing at the output pin gives a maximum frequency of 259 MHz. This is the maximum frequency where the –1-dB compression would be 17 dBm at the matched load. Higher useable bandwidths are possible at lower output powers, as shown in the Large Signal Bandwidth curves. As those graphs show, 7-V_PP outputs are possible with almost perfect frequency response flatness through 100 MHz for both non-inverting or inverting operation.

8.3.7.3 Noise Figure

All fixed-gain RF amplifiers show a very good noise figure (typically < 5 dB). For broadband amplifiers, this is achieved by a low-noise input transistor and an input match set by feedback. This feedback greatly reduces the noise figure for fixed-gain RF amplifiers, but also makes the input match dependent on the load and the output match dependent on the source impedance at the input.

The noise figure for an operational amplifier is always higher than for fixed-gain RF amplifiers, due to the more complex internal circuits of an operational amplifier (giving higher input noise voltage and current terms). Also, for simple circuits, the input match is set resistively. What is gained is an almost perfect I/O impedance match, much better load isolation, and very high 3rd-order intercepts versus quiescent power. These higher noise figures can be acceptable if the OPA695 has enough gain preceding it in the IF chain.

Operational amplifier noise figure equations include at least six terms (see Noise Performance), due to the external resistors. As a point of reference, the circuit of Figure 48 has an input noise figure of 14 dB, while the inverting configuration of Figure 49 has an input noise figure of 11 dB. At higher gains, it is typical for the inverting noise figure to be slightly better than for an equivalent gain, noninverting configuration. Improve the noise figure for the noninverting configuration of the OPA695 by including a step-up, 1:2 turns ratio transformer at the input. This configuration is shown in Figure 52.

Figure 52. IF Amplifier With Improved Noise Figure

The transformer provides a noiseless voltage gain at the expense of higher source impedance for the OPA695 noninverting input current noise. The input impedance is still set to 50 Ω by the 200-Ω resistor on the transformer secondary. A 1:2 turns ratio transformer will reflect the 200 Ω to the input side as a 50-Ω impedance over the bandwidth of the transformer. Using a 1:2 step-up transformer also reduces the required amplifier gain by 1/2 for any particular desired overall gain.

Table 1, Table 2, and Table 3 summarize the recommended resistor values and resulting noise figures over the desired gain setting for three circuit options for the OPA695 operated as a precision IF amplifier. In each case, R_F and R_G are adjusted for both best bandwidth and required gain.

In all cases, exact computed values for resistors are shown; in an application, pick standard resistor values that are closest to those in the tables.

Table 1. Noninverting Wideband Operational Amplifier

GAIN TO LOAD (dB)	RF (Ω)	RG (Ω)	NOISE FIGURE
6	478	159	17.20
7	468	134	16.55
8	458	113	15.95
9	446	96	15.40
10	433	81	14.91
11	419	68	14.47
12	402	57	14.09
13	384	48	13.76
14	363	40	13.23
15	340	33	13.23
16	314	27	13.03
17	284	21	12.86
18	252	16	12.72
19	215	12	12.60
20	174	9	12.51

Table 2. Noninverting With a 1:2 Input Step-Up Transformer

GAIN TO LOAD (dB)	RF (Ω)	RG (Ω)	NOISE FIGURE
6	516	518	16.34
7	511	412	15.54
8	506	334	14.78
9	500	275	14.07
10	493	228	13.40
11	486	190	12.78
12	478	160	12.21
13	469	135	11.70
14	458	114	11.25
15	447	96	10.85
16	434	81	10.15
17	419	69	10.21
18	403	58	9.96
19	384	48	9.74
20	364	40	9.57

Table 3. Inverting Wideband RF Amplifier

GAIN TO LOAD (dB)	Optimum RF (Ω)	RG (Ω)	Input Match RT	NOISE FIGURE
6	463.27	116	87	16.94
7	454.61	101	98	16.06
8	444.91	88	114	15.16
9	434.07	77	142	14.23
10	421.95	66	199	13.24
11	408.42	57	380	12.16
12	398.11	50	Infinite	11.03
13	446.68	50	Infinite	10.92
14	501.19	50	Infinite	10.83
15	562.34	50	Infinite	10.75
16	630.96	50	Infinite	10.67
17	707.95	50	Infinite	10.61
18	794.33	50	Infinite	10.55
19	891.25	50	Infinite	10.49
20	1000.00	50	Infinite	10.45